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l Copr. 1949-1998 Hewlett-Packard Co. 

A New High-Speed Multifunction DVM 

Plug-ins provide true rms ac capability as well as dc 
and ohms. Reading speed is 1000 per second of ohms 

and dc. 

By Craig Walter, H. Mac Juneau and Lee Thompson 

there is A need today for 'horizontal expansion' of the 
capability to measure dc and ac voltage, and resistance — 
more accuracy in general applications, with good repeat- 
ability. In addition, there is a need to reduce the difficulty 
of eliminating errors under conditions such as making a 
floating dc measurement in the presence of both common 
mode and normal mode error signals, or avoiding large 
errors when measuring distorted sinusoids or waveforms 
without zero axis symmetry. 

For bench users, the need is not to make an already 
difficult measurement with greater precision, but there 
is a need to make measurements of adequate resolution 
with more ease and reliability. The measurement prob- 
lems of the bench user have often been ignored in favor 
of 'greater' or 'more' rather than 'better'. Recently some 
instruments have been compromised in favor of the 'sys- 
tem' aura — instrument optimization for system use at 
the expense of, rather than for. the bench user. 

The system user can generally find a unique solution 
to his unique problem. Having found it. he can operate 
his system properly. He has the time and generally the 
capital to find unique solutions to his individual prob- 

The bench user has a difficult problem each time he 
uses the instrument. A lash-up that provides optimum 
results for a measurement one day cannot be expected 
to yield the same results if the measurement problem 
changes. The bench user's measurement problems vary 
from day to day. and he seldom has time or money to 
invent solutions for each problem. 

Nevertheless, the problems associated with instrument 
use in both applications are not totally unique to either. 
A great degree of commonality exists. Providing solu- 
tions for the common measurement errors that exist in 
bench applications can result in an instrument useful 


for systems use at little or small expense to the bench user. 

Indeed, the primary distinction between the two appli- 
cation areas often is in the speed ( and end use) of the 
measurement, not in the measurement itself. The instru- 
ment, in bench applications, is interfaced to a human; 
in system applications, to a machine. If the same meas- 
urement can be made with greater speed (and if. in addi- 
tion, the data collected is easily transferred to a machine 
as well as a human), the instrument can also satisfy many 
system requirements. It should be possible, then, to pro- 
vide an instrument that could properly be called a hybrid 
— optimized for bench and widely useful in systems. 


l Copr. 1949-1998 Hewlett-Packard Co. 

Cover: Both half-module 
and rack versions of the 
Hewlett-Packard Model 
3480A/B are shown. Its 
reading speed is 1000 per 
second for ohms and dc to 
1000 volts. True rms ac is an 
option. This article dis- 
cusses dc processing and 
preconditioning as related to the Model 3480 A/ B. 
The ac conversion technique will be covered in 
detail in a future issue of the Hewlett-Packard 

In this Issue: 

A New High-Speed Multifunction 
DVM, by Craig Walter, H. Mac 
Juneau and Lee Thompson P°S e 2 

Fig. |, This new Hewlett-Packard Model 3480A/B is a high speed multi-tunction DVM 
capable ol making 1000 readings per second up to WOO volts dc, and ohms down to 100 
ohms lull scale. It comes in both hall module and rack versions. Also shown here are 
the Models 3481 A Butler Amplitier (with one 10 V dc range), 3482A DC Range Unit and 

the 3484A Multifunction Unit. 

Bench Features 

Many features of the new HP Model 3480A/B, Fig. 1, 
which may seem mundane by themselves, combine to 
make the instrument easy to use. The display is easily 
readable, function and range information and the instru- 
ment's functions are readily apparent. Autoranging is 
complete through all ranges and functions. The first read- 
ing after an autorange cycle will be correct. The sample 
rate is fully controllable, from a sample initiated by a 
front panel pushbutton to >25 readings per second; 
higher sample rates — to 1000 per second — can be initi- 
ated by external commands. Selectable filtering is pro- 
vided for normal mode signals so that the user has a 
variety of choices between instrument settling time and 
interference rejection. The instrument is fully protected 
from damage from overvoltage. on any range, in any 
function. Overload recovery of the amplifiers is fast 

enough that a correct reading upon removal of the over- 
load is guaranteed, even though the reading is started at 
the same time the overvoltage is removed. 

Zeroing requirements are held to a minimum. The 
instrument has accuracy commensurate with its resolu- 
tion — at moderate or extreme speeds. The ac converter 
uses a thermopile to provide true rms conversion to elimi- 
nate common ac measurement errors. And of great im- 
portance, the instrument has minimum effect on the 
device or circuit under test — all injected currents have 
been eliminated or reduced to an insignificant level. 

Measurement Speed 

Terminology usually undergoes transformation when 
applications are changed; hence, for system use, the 
DVM is more properly called an A/D (analog to digital) 
converter. There can be significant differences. 

© Copr. 1949-1998 Hewlett-Packard Co. 

For bench applications, the sequence of readings per 
second should be quick compared with human reaction 
time. A good performance criterion is the time required 
for a single measurement, limited by the ability of the 
analog circuits to respond to sudden input changes and 
settle to the final value. 

Most systems are without similar restrictions — data 
can be absorbed at much faster rates. Thus, high reading 
speed is a primary concern, provided, of course, that 
response and settling times within the instrument are 
commensurate with its reading rate. 

Often speed is a major distinction. Because of the 
design compromises generally required to obtain high 
system speeds the system-designed analog to digital con- 
verter (A/D) is ordinarily less versatile than its bench 
counterpart. It is task dedicated. A DVM, although slow- 
er, has varied functional capability and more versatile 
signal preconditioning. Does the versatility necessary 
for bench use preclude the measurement speed deemed 
necessary for systems applications? 

In many system applications, the Model 3480A repre- 
sents a solution to this apparent paradox. For systems 
use it can be considered a comparatively slow A/D (its 
maximum sampling speed is 1000/s) with 15 bit resolu- 
tion and much greater signal-conditioning capability than 
the typical A/D. As such, it may be the best candidate 
for certain system situations. 

The digitizing technique used — successive approxima- 
tion — provides moderate speed at low cost. It is inher- 
ently simple and quite reliable. Reed relays in the A/D 
converter are replaced with semiconductor switches. The 
instrument accommodates plug-ins to provide signal pre- 
conditioning (giving great measurement versatility), and 
the main frame contains the necessary power supplies and 
the A/D. The performance of either section is comple- 
mented, not compromised, by the other. 

To avoid placing an undesirable burden on the bench 
user — added costs without benefit — the interfacing cir- 
cuitry required to communicate with other instrumenta- 
tion is not included within the basic instrument. These 
interfacing circuits are. instead, available as options. 

The emphasis during development was to capitalize on 
the digitizing speed made available by the successive 
approximation technique without compromising bench 
performance or increasing the instrument's basic cost; to 
solve those problems common to all traditional measure- 
ments, not those related to instrument use in a single 
application — bench or system. 
Measurement Errors 

Errors associated with instrument use commonly fall 

within three groups. One group is caused by the measure- 
ment circuit's interaction with its surroundings. The most 
common sources are normal mode and common mode 
generators which, directly or by magnetic coupling, in- 
duce unwanted currents in the measuring loop; the 
sources can be magnetic fields from other instrumentation 
or voltages generated because of the flow of relatively 
large currents in the ground connections among instru- 
ments. A second error group is caused by interactions 
between the measuring instrument and the circuit or de- 
vice under test. Common mode or normal mode sources 
exist within virtually all instrumentation, and may 
force 'injected currents' into the circuits being measured; 
this may occur between 'high' and 'low' or between the 
instrument's chassis and its other input terminals. These 
currents can create errors by flowing through unbalanced 
impedances in the input circuit or, more importantly, may 
actually upset or change the characteristics of the circuit 
under test. Third among error sources is those caused 
directly by the instrument. The most obvious are errors 
in amplification, attenuation, or conversion that somehow 
modify the information being sought so that the data 
presented as absolute may actually be in error. 

Unfortunately, reducing the errors of one group does 
not guarantee reduction of the others. In fact, the opposite 
is often true. These sources of error, although classified 
separately, must be treated simultaneously to minimize 
the entire error matrix. 


CM and NM Sources— Outside the DVM 

The most general measurement situation is shown in 
Fig. 2. A floating (above earth or chassis ground ) meas- 
urement is to be made across a resistance bridge. Al- 
though the instrument ground and the source ground are 
on the same line, a voltage generator (the common mode 
source) will exist between them. The difference in the 
ground voltage is primarily due to induced currents and 
ground currents that flow between the two physically 
isolated grounding points. This current creates a voltage 
difference (the ground line or plane will always have 
some impedance) whose magnitude depends on the hook- 
up and the environment into which it is placed. 

The common mode generators (ac and dc) will not. 
by themselves, cause measurement errors if the instru- 
ment impedance from chassis to each of its input termi- 
nals is infinite. Unfortunately, this is not generally the 


© Copr. 1949-1998 Hewlett-Packard Co. 

Fig. 2. A general measurement situation where a tloating 
measurement is made across a resistance bridge. Its 
Th6venin equivalent is shown. 

Fig. 3. Possible common mode currents from external 
sources. Assume Z. < < Z, and Z,. 

case. Because impedance is finite between the instru- 
ment's high and low terminals and the point to which the 
common-mode generator is referenced (the instrument's 
chassis), Fig. 3, current will flow through each of the 
imbalance resistors. The resulting voltage drop across 
these imbalance R's will enhance or oppose E, r ., creating 

normal mode errors — the common-mode signal has been 
converted to one in series with the primary measurement 
loop, that is, to a normal mode error signal. (The treat- 
ment of common mode errors, once the conversion to 
normal mode occurs is then identical to that for normal 
mode errors.) These signals create undesirable errors — 
either dc offsets or a time varying voltage which may 
cause the DVM's display to 'rack.' 

Because the instrument's 'high' terminal is generally a 
point or line rather than a plane, its impedance to chassis 
is generally quite high and common mode errors in this 
input lead can generally be ignored. If, however, the im- 
pedance from high to low (>10' u n, <50 pF for each of 
the 3480's plug-ins) is not extremely high and a guard is 
either not available or used, significant errors can result 
— a 60 Hz common mode voltage of 10 V, an imbalance 
R of 1 left, and an input capacity of 1 000 pF combine to 
generate an error of 3.7 mV. Normal mode filtering can, 
of course, reduce the ac errors but not without a corre- 
sponding increase in measurement time (the response 
time of the filter must be added to the instrument's basic 
digitizing time). 

The instrument impedance from low to chassis is 
usually much lower because of the instrument's physical 
construction; 'low' will generally be a plane — capacities 
of several thousand picofarads are not unusual. Guarding 
must then be used to reduce these errors. Connecting the 
guard. Fig. 4, will effectively bootstrap that portion of the 
impedance between low and chassis that is terminated on 
guard. It can be of little help, obviously, for that imped- 
ance not interrupted by guard. 

Fig. 4. Proper guard connections will shunt most com- 
mon mode current away trom source resistances. 


© Copr. 1949-1998 Hewlett-Packard Co. 

The degradation in Z„ that is so undesirable ean easily 
occur when the instrument is interfaced to other equip- 
ment. If. for example, the data generated by the DVM is 
to be used to provide hard copy, a printer or other record- 
ing device will be electrically tied to the DVM's data 
output lines. If. too, the instrument is to be remotely 
programmed or externally commanded, the program 
source must be electrically connected to the DVM. As 
the DVM's data programming lines are referenced to the 
instrument's low terminal, a floating measurement will 
be impossible unless this added instrumentation can also 
be floated. Floating this instrumentation, unfortunately, 
will reduce Z„ and cause deterioration of the system's 
CMR. Injected currents from low to chassis will also be 
significantly increased — unless the output or controlling 
circuitry is completely isolated from its chassis. If the 
output and control lines can be so referenced, and if iso- 
lation can be provided within the DVM, these system 
errors can be eliminated. The Model 3480A/B and its 
plug-ins have digital output and programming options 
for necessary electrical isolation without degrading other 
performance criteria ( measurement speed, susceptibility 
to electrical interference). The isolated programming op- 
tion also provides program storage. 

The physical architecture of the Model 3480A/B has 
eliminated the necessity for costly 'box-within-a-box-con- 
struction' fall internal circuitry surrounded by guard) — 
yet CMR for the Model 3480A/B and any of its plug-ins 
is >80 dB at 60 Hz for a 1 kn imbalance. Physical spac- 
ing between the internal circuitry ('low') and chassis is as 
large as practicable. Where large spacings are impractical, 
individual shields are employed. 'Box-within-a-box' con- 
struction is necessary only when much higher levels of 
CMR arc required. While reducing errors caused by ex- 
ternal CM voltages, this type of construction may accen- 
tuate measurement errors caused by injected currents. 
Where guarding is required within the instrument (the 
transformer, power supply heat sink, and plug-in covers 
are the principal guard shields) the necessary care has 
been taken to eliminate time-varying voltages in their 
vicinity. The result is an extraordinarily favorable set of 
tradeoffs. The complete measurement problem has been 
taken into account. Errors from all sources, not just a few, 
have been reduced together in appropriate amounts. 

Normal Mode Filtering 

Normal mode filtering, again, can be used to reduce 
these errors, but measurement speed is reduced. The most 
obvious solution to this tradeoff is to use a digitizing 
technique that provides filtering — i.e., integration. Hence. 

the recent popularity of dual-slope DVM's. This com- 
promise does not solve the entire measurement problem 
unless the injected currents are also minimized. Those 
currents may do nothing to the instrument because of its 
inherent rejection, but can and do create subtle and seem- 
ingly mysterious changes in the circuit under test. Inte- 
gration is also only effective for noise whose frequencies 
are related to (and multiples of) the converter's integration 
period. Although filtering by integration can, theoreti- 
cally, give superior results at these discrete frequencies. 
Fig. 5, little help is afforded the user whose noise is not 
exactly synchronized to the DVM's integration period. 
Moreover, an ideal converter using integration is limited 
to a maximum sample rate of 60 Hz (and a correspond- 
ing minimum aperture time of 16.6 ms) if the CM fre- 
quency is exactly 60 Hz. 

Fig. 5. Normal mode rejection at discrete frequencies 
characteristic of a typical integrating DVM 

Normal mode rejection can also be achieved by passive 
or active filtering or by a combination of filtering and 
frequency conversion. A chopper stabilized amplifier is a 
good example. Filtering may occur before, within or after 
this input amplifier. All have relative advantages and dis- 
advantages — none is an ideal solution to the general 
measurement problem. The degree of filtering required 
for different applications will, of course, be different as 
will the measurement times desired. There will always be 
a speed/ rejection tradeoff, stated or implied. Rather than 
restrict the user to a fixed compromise — the amount of 
filtering and the delays that are a necessary consequence 
— filtering in the plug-ins for the Model 3480A/B is 
selectable (see Specifications). The user can choose the 

© Copr. 1949-1998 Hewlett-Packard Co. 

compromise that best suits his needs. 

Strong magnetic fields near the DVM can contribute 
both common mode and normal mode errors if care is not 
exercised in the design of the instrument. Here, filtering 
may be of no direct benefit, for the injected currents may 
be induced in the instrument's filter or in the circuitry 
following the filter or the integrator (if the DVM uses 
integration). A five gauss 60 Hz field (typically found 
near the primary power section of most instrumentation) 
can induce a peak-to-peak error as large as 30 /<V if the 
circuitry within the field encloses an area of one square 
inch. Loops can be generated by the improper layout or 
design of either or both of the DVM's high and low leads. 
Shielding will be of little help if the field is large enough 
to cause saturation. 

Reducing these errors within the Model 3480A/B and 
its plug-ins is accomplished in several ways. All the input 
wiring — -high and low — form tightly twisted pairs to re- 
duce the area that may enclose the field. Where twisted 
pairs are impractical, compensating loops have been 
added. Wirewound resistors, notorious for their ability to 
sense magnetic fields, even when 'non-inductively* wound, 
have been replaced in sensitive areas by precision metal 
film resistors. 

CM and NM Sources Inside the DVM 

Not all common mode currents (and the normal mode 
errors they create) come from the circuit being measured. 
Some common mode error sources are generated within 
the measuring instrument. These are caused by currents 
induced into the ground or guard shields by voltages 
referenced to chassis, or into chassis from sources refer- 
enced to low. These internal common mode sources are 
generally constant current sources and force 'injected 
currents' into the circuits being measured, Fig. 6. Direct 
measurement errors are the normal result, but by upset- 
ting the circuit under test or by changing its characteristics 
during the measurement period, indirect errors often oc- 
cur — these errors, because they occur only when the 
device under test is connected to the DVM. are often 
impossible to isolate and identify. 

The most common source of these injected currents is 
the instrument's power transformer. The transformer, to 
reduce its capacity from low (secondary) to chassis (pri- 
mary ) is guarded. Capacity from chassis to low is inter- 
rupted by a guard shield between the two windings. Fig. 
7. If the guarding is complete (C, quite small) the injected 
current flowing through low will be correspondingly small 
(if E, is 100 V at 60 Hz and C, = 10 pF, the result cur- 
rent — 400 nA — will develop 400 /iV across an R b of 1 k. 
Fig. 6). 

Fig 6 Internally generated common mode sources rel- 
erenced to chassis. 

The injected current flowing through the guard termi- 
nal — caused by C. — is of no consequence as it shunts the 
measurement circuit. If. however, the guard is connected 
to low. all the injected current will flow through the meas- 
urement circuit. This error can be hundreds or thousands 
of times larger than the one previously calculated (C ; 
1000 pF). This error source can be reduced by placing 
an additional shield between the existing guard shield and 
the transformer primary winding. 

Other sources of similar magnitudes exist between the 
transformer's secondary winding and its guard shield, 
Figs. 7 and 8. These inject current from low to guard. 
This injected current will only go through R b if the guard 
is properly connected; if tied to low the current is shunted 
around the measurement circuit. So connecting the guard 
to maximize CMR will also maximize the effect of this 
particular injected current. Connecting it to minimize the 
injected current will correspondingly reduce the CMR. 
The need for this tradeoff can be eliminated if still 
another shield — between guard and low — is added to the 
transformer. Fig. 9. 

The transformer construction in the Model 3480A/B 
incorporates all three shields to reduce all of these injected 
currents. Both the primary and secondary windings are 
completely enclosed in 'box' shields tied to their respec- 
tive grounds. A guard shield between these two boxed 
windings is used as a guard to maximize CMR. In this 
manner injected currents are limited to a few nanoamps. 

Shielding requirements for the transformer must also 
extend to all primary and secondary wiring — in fact to all 
time varying voltages within the DVM. The primary wir- 


© Copr. 1949-1998 Hewlett-Packard Co. 

Fig. 7. A typical guarded transformer and capacitances 

ing is physically isolated and shielded from circuitry 
referenced to low. The secondary wiring and all the recti- 
fication and regulation circuitry used for the DVM's in- 
ternal power supplies are also shielded and isolated from 
chassis (the instrument's frame). 

Timing, gating, and external display circuitry must 
also be shielded (and guarded, if necessary) from the 
instrument chassis. Logic circuitry that generates internal 
timing and gating is located on a single printcd-circuit 
card in the middle of the instrument — boards on either 
side shield it from chassis. The sample rate generator that 
initiates each sample is coupled to the plug-in by a steady 
state voltage, not one that is time varying. Time varying 
voltages on the 'mother board' are shielded and guarded 
from the chassis by another board below. This board, 
physically attached to the mother board, also provides the 
mechanical stiffening necessary to insert and extract the 
other plug-in cards from the mother board. 

Gas discharge display tubes are also isolated. Because 
of the differences in the glow area of the various segments 
(glow tube cathodes), each has a different sustaining 
voltage. Because of the differences in the spatial arrange- 
ment of the segments, every unlit segment assumes a 
unique voltage — a voltage that will be dependent on the 
lighted segment. These voltage variations can be in excess 
of 40 V. When the display is changed, these voltages also 
change, and create large voltage transients that can gen- 
erate large injected currents (the capacity between the 
Nixie segments and the instrument's chassis is relatively 
large because of the glow tube's large surface area). Isola- 
tion is achieved by depositing a metallic coating (tied to 

Fig. 8. Internally generated common mode sources ref- 
erenced to 'low.' 

low) on the inside of the plastic window that is the front 
of the mainframe. This conformal coating, although it 
does provide significant attenuation of the broadband 
noise generated by these display tubes, is not sufficient to 
reduce the injected currents to the nanoamp levels de- 
sired. Therefore, buffering between the decoder drivers 
and the D/A logic has been added. The display is changed 
only once — at the completion of each reading. From a 
visual standpoint, buffering is not required because of the 
relatively small digitizing time of the A/D — the human 
eye could not detect the change in the voltage displayed 
during digitizing. 

Shielding has also been added to the board on which 
the D/A converter and the comparator are located. 
Shielding is required here because the physical spacing 
between the components on the board and top cover is 
not sufficient to reduce the injected currents from these 
sources to acceptable levels. These components are delib- 
erately near the top of the instrument to provide easy 
accessibility to calibration potentiometers. The heat sink 
on the regulator card must of necessity, be tied to guard 
(its capacity to the instrument top cover cannot otherwise 
be tolerated). Here again, all time varying circuitry refer- 
enced to low has been physically removed from the vicin- 
ity of this shield to reduce injected currents. Time varying 
lines between the mainframe and plug-in are either 
shielded (and guarded, as necessary) or have voltage 
levels reduced commensurate with the injection desired. 

This method of guarding and shielding to minimize 
injected currents without the necessity of sacrificing CMR 
in actual use has also been used in the plug-ins. the iso- 


© Copr. 1949-1998 Hewlett-Packard Co. 

Fig. 9. Manner in which the transformer in the Model 
3480A is shielded and guarded. 

lated digital output option (for the mainframe), and the 
isolated programming option (for the plug-ins). The total 
injected current from all sources has been reduced to a 
few nanoamps — a level sufficient to eliminate any error 
when moderate unbalances are used, regardless of the 
guard connection. 

Current injection from the instrument high terminal to 
low will also cause an obvious error. Leakage or injected 
currents are dc rather than time varying (currents used 
to bias the input amplifier or currents from improperly 
shielded power supply voltages). The total input error 
involves not only the instrument's input resistance but 
also this dc leakage current. An instrument with lO'-n 
input resistance may have a leakage current of 1 nano- 
amp. If a source resistance of 1 MO is used, little loading 
error will result, but the offset caused by the leakage 
current will be 1 millivolt. The plug-ins for the Model 
3480 A/B have an initial offset current of < 10 picoamps; 
its change with temperature (perhaps of more impor- 
tance) is less than 1 picoamp/°C. 

Input impedance (and offset current) of the instrument 
is also constant with time or sample rate. 'Kickback' cur- 
rents that might adversely affect measurements from rela- 
tively high source impedances have been eliminated. 


DC Signal Preconditioning 

Conditioning the signal voltage to the nominal value 
required by the A/D ( 10 V) within the accuracies desired 
(±0.005%) can be achieved easily. It is more difficult 

though, if characteristics other than accurate amplifica- 
tion or attenuation are also required. Some of the more 
obvious requirements are: moderate bandwidth (20 kHz 
(u A = 40 dB); wide dynamic range (0 V to ± 15 V at 
the instrument's input); extremely high input resistance 
( > 10"'O); very low offset voltage and current (< 1 /iV, 
1 pA at the amplifier input); and low sensitivity to power 
supplies, source and load impedances, temperature and 

All but the requirement for bandwidth are associated 
with the design of any low level dc amplifier. The actual 
operating characteristics desired are fast recovery from 
overload (>50 ,us) and a slew rate and settling time fast 
enough to make useful the A/D digitizing time. These 
requirements imply a bandwidth in excess of 20 kHz. To 
satisfy the bandwidth requirement only is simple. But to 
satisfy both the requirement for dc preconditioning and 
for bandwidth requires greater sophistication. 

Chopper amplifiers (which up-convert the signal to 
some low carrier frequency, amplify, then down-convert) 
will not normally satisfy the bandwidth requirement. Such 
amplifiers are normally used only to amplify dc and low- 
frequency voltages near dc. To amplify the higher fre- 
quencies, an ac-coupled amplifier can be paralleled, but 
this is. of course, more complicated and more costly. 
Up-conversion to a much higher frequency carrier (mega- 
hertz), as in a parametric amplifier, accommodates the 
frequency range requirement, but adequate amplifier ac- 
curacy and dc stability (variation of the offset voltage 
and current at the amplifier input with time and tempera- 
ture) are difficult to achieve. 

A direct-coupled amplifier of unusual design was the 
final choice for the instrument, satisfying performance 
requirements at reasonable cost. 

A matched pair of field-effect transistors is used for the 
differential input stage, Fig. 10. The FET offers both the 
high input resistance and the small leakage current re- 
quired. Bipolar devices, though not necessarily limited 
by their lower input resistance (although lower than the 
FET. it is boosted by the amplifier's loop gain) have con- 
siderably more leakage current. The FETs are operated 
in a balanced common drain configuration to achieve 
minimum sensitivities of offset voltage and gain to varia- 
tions in power supply and device parameters. High CMR, 
>80 dB, is not readily obtainable in a common source 
configuration because of the inherent mismatch of the 
two discrete devices. 

To reduce parameter variations caused by temperature 
fluctuations, the FET environment is temperature con- 
trolled at a temperature higher than the maximum ex- 

© Copr. 1949-1998 Hewlett-Packard Co. 

pccted ambient, An integrated circuit is used as an 'oven' 
to maintain the FET at constant temperature. The mono- 
lithic IC has within it all the circuitry normally associated 
with an oven and its control circuitry — heaters, tem- 
perature sensors, and amplification. The FET dice, 
mounted atop the IC. assume the temperature of the 
larger chip. Although the temperature control does oper- 
ate open loop (the sensing devices are within the IC, not 
the FETs), the resultant thermal gain (-\V« ;s without 
temperature control/ A V (1S with temperature control) can 
be quite high (A T > 100). A high thermal gain, however, 
does not guarantee a reduced offset voltage temperature 
coefficient. The thermal gains of the two devices must be 
matched because of their large initial TC ( ~ 600 ».V/°C), 
i.e.. if the two devices are ideally matched initially but 
A T = 100 for one side and 1000 for the other, the net TC 
will be 5.4 ;<V/°C, not zero. Compensation must be used 
to reduce the effects of open loop control. 

The device is constructed with obvious symmetry 
about the two FET dice. Fig. I 1 . Thermal gradients 
across the face of the IC have been reduced by an 
anodized aluminum heat sink (0.001" X 0.030") be- 
tween the IC and the FET chips. The epoxy used for 
mounting the FETs and the IC is thermally conductive 
although electrically resistive. The aluminum bonding 
wires (1.5 mil diameter) used for connecting the FETs 
are thermally bootstrapped. Such bootstrapping is re- 
quired to minimize the effect of the heat conduction 
through the bonding wires — >60% of the heat lost. If 

Fig. 10. Input stage ot the 
Model 3482A and Model 3484A 
dc preconditioning amplifier. 

heat is lost unevenly, the gradients that result are severe 
enough to seriously degrade both the absolute value of 
thermal gain achieved and the resulting match in thermal 
gain between devices. 

The FETs are operated at 80°C, a temperature high 
enough above ambient to allow regulation when the 
instrument is operated at elevated temperatures. Com- 
pensation reduces the offset current initially to < 1 0 pA 
and attains a composite TC of <1 pA/°C. To keep the 
resulting, offset current independent of input voltage level, 
the compensation circuitry is bootstrapped. 

Even though the temperature of the FET is controlled, 
the offset voltage temperature coefficient of the FET, 
combined with the rest of the amplifier, may still be 
greater than desired. To reduce this to <±1 ^V/°C the 
entire amplifier is temperature compensated. The ampli- 
fier's temperature is varied and its TC calculated. 
Resistive compensation is added, that is the amplifier's 
TC is changed and the amplifier is rerun until the desired 
TC is achieved. Compensation is achieved by varying the 
V BE match of the first bipolar gain stage. Its TC match 
will change by approximately 1 /iV/°C for each 300 jtV 
mismatch in its base-emitter voltage.'" Resistors are 
added in scries with the emitters of these transistors to get 
the required mismatch. As the stage is driven by a current 
source, the voltage drops across the emitter resistors act 
as small batteries to mismatch AV BE . 

To take full advantage of the DVM's reading speed, 
field effect transistors are used for dc range switching. 

© Copr. 1949-1998 Hewlett-Packard Co. 

FET Chips 



AL Strip 

Fig, II. Construction ol the FETs on a chip with temperature compensation. Heaters 

are on the same chip. 

Input voltages from 10 to 1000 V are divided down to 
10 V by reed relays. Ranging is then by FETs. The FET 
*on' resistance, although several hundred ohms, is in 
series with the amplifier's input impedance f > 10-'5J) and 
thus creates little error. The leakage current from the 
inverting side of the amplifier (and from the 'off switches) 
flows through the "on' switch and. on the two lower 
ranges. 10 k'.}. Fig. 12. Even though this leakage current 
can be as large as 1 nA. it is relatively constant because 
of the controlled environment of the FET used as the 
amplifier input stage. The offset it creates can be removed 
initially by zeroing. Any changes with temperature are 
accounted for by the compensation technique previously 

Fig. 12. DC amplifier gain switching. 


© Copr. 1949-1998 Hewlett-Packard Co. 

What is the 
HP Model 3480 A? 

The Model 3480A/B is a 4-digit digital voltmeter (with 
50% overrange). an A/D converter with moderate speed 
which, when combined with one ot its plug-ins, can provide 
multifunction measurement capability without many of the 
limitations created by traditional measurement errors. 

The 3480A/B mainframe (the 'A' is a half module; the 
'B'. a full rack module) uses plug-ins — Models 3481A, 3482A 
or 3484A. The Model 3484A, with all options, has five dc and 
true rms ac voltage ranges, and six ohms ranges. The Model 
3482A has the same dc capability as the Model 3484A (it 
cannot, however, be expanded to provide ac and ohms). The 
Model 3481A has only a single dc voltage range. All plug-ins 
fit either mainframe configuration. 

Successive approximation is used for A/D conversion. 
Because of the design of the analog processing portions of 
the instrument (within the plug-in) and the means employed 
for data or programming transfer, reading and recording 
speeds up to 1000 per second are possible without per- 
formance degradation. 

A true rms ac converter (an option within the 3484A) en- 
ables accurate voltage measurements to be made of wave- 
forms with frequency components from dc to 10 MHz. The 
converter eliminates significant errors (when other conver- 
sion techniques are used) caused by small amounts of har- 
monic distortion present in most sinusoidal signals. Accurate 
measurement of non-sinusoidal phenomena is also possible 
— the full scale crest factor is 7:1. Because the converter 
can be dc coupled, it also measures the rms value of a com- 
bined ac and dc signal. A dual-thermopile makes the con- 
version and is 30 times more sensitive than a single thermo- 
couple. This sensitivity permits accurate measurements on 
the 100 millivolt range. 

DC input errors between the high and low input terminals 
are virtually eliminated by the combination of a constant 
input resistance of >10 : ohms and a leakage current of 
<10 pA. A three position input filter can be used to reduce 
or eliminate measurement errors caused by normal-mode 
noise. Errors caused by common-mode noise can be re- 
duced by using the guard. Injected currents flowing from 
the low and guard terminals to chassis have been signifi- 
cantly reduced to minimize the effect the instrument has 
on the device or circuit under test. 

System options include isolated or non-isolated BCD out- 
puts and isolated programming inputs. Everything (except 
terminal selection) on the DVM is programmable. A two 
range, three terminal, dc ratio option is also available. The 
variety of possible configurations available and the innate 
operating features allow the user to easily adapt the instru- 
ment to his specific needs while simultaneously reducing 
measurement errors. 

Fig. 1 3. FET range switch. 

Ql and its associated components drive 02 'on' or 
'off'. Fig. 13. When the range line is open (high), 01 
is off and the gate is biased to the negative supply through 
R4 and CR 1 . Grounding the range line reverses biases 
CR1 — Q2 turns on and is zero biased through R3. 


Paul Baird and Ken Jessen contributed significantly to 
the project's definition. George Latham was responsible 
for the 3480A/B's electrical design. Dave Luttropp for 
its mechanical design. Jim Arnold's suggestions greatly 
increased the instrument's serviceability. John Hettrick 
bore the primary responsibility for coordination of the 
production transfer, Gregg Boxleitner the responsibility 
for production and electrical design of the 3481 A. Karl 
Waltz and Barry Taylor were jointly responsible for the 
3482A, Mike Aken for the 3484A. Jerry Blanz was 
responsible for the mechanical design of the plug-ins. 
Approbation must also be given to Larry Lopp and Jerry 
Harmon who contributed to the SFET's development and 
to Larry Linn who made many significant contributions. 
Too numerous to name but nonetheless indispensable are 
the many other people involved in layout and manufac- 
turing. " 


1 A. H. Hoffait and R. D. Thornton. "Limitations of Tran- 
sistor DC Amplifiers." Proceedings of the IEEE. February 


© Copr. 1949-1998 Hewlett-Packard Co. 

Electrical Isolation: Coupling from Low to Chassis 

Transformer coupling is used to transfer the information 
used for programming or digital output — from circuitry ref- 
erenced to low, to or from circuitry referenced to chassis. 
This solution offers both speed and reliability — neither 
achievable with reed relays. It has one inherent disadvan- 
tage, however. The successive approximation technique, 
unlike integration, presents the data in a parallel, rather 
than serial format. The information to be transferred is in 
its final form. Data transfer, then, requires a separate trans- 
former for each bit — 32 for data, 15 for programming. 

Integrating or digitizing techniques that use voltage to 
frequency converters need provide transfer only for their 
clock pulses (a single line). Decoding is then done in the 
'out guard,' or chassis section of the instrument. Program- 
ming isolation is inherent in the reed relays typically used, 
so additional transfer is not necessary. 

The costs involved in implementing a multiple transformer 
scheme at first appear prohibitive. But to convert first from 
the parallel format to serial, transfer, and then reconvert is 
also costly. Serial transfer also requires clocking to main- 
tain cogency. At the speeds deemed necessary for data 
transfer, the injected currents caused were considered ex- 
cessive. The use of light isolators, although attractive, is 
still somewhat expensive. 

To reduce costs, the transformers used are simply pairs 
of molded RF chokes. DC isolation is achieved by mounting 
the chokes and their respective circuits on interfacing PC 
boards — one referenced to low and one to chassis. Power 
for the isolated or chassis side is provided by a separate 
winding on the transformer and by a separate regulator — 
both isolated from the instrument's ground (low). 

The basic coupling circuit is shown below. 

these common mode voltages may be switched at rapid 
rates (when used with a scanner switching large high, low, 
and guard voltages), the shielding and the guarding it im- 
plies are necessary. 

A small capacity (<1 pF) exists between each of the two 
coils used for transfer. In the example shown, a large and 
relatively fast common mode voltage (caused by switching) 
may inject enough current Into the base of the transistor 
to cause the latch to change state. This injected current — 
caused by the external circuitry, not by the DVM— can be 
eliminated if a shield, tied to chassis, is used to interrupt 
the capacity between coils. If information is to be trans- 
ferred in the opposite direction, the shield must be tied to 
low. Unless these shields are properly terminated, current 
injection will be enhanced and noise sensitivity reduced. 

These shields, although providing the reduced sensitivi- 
ties desired, greatly decrease CMFt. An additional shield, 
tied to guard, is added to obtain a net reduction in capacity. 
A compromise between CMR and noise sensitivity is re- 
quired, however, as common mode voltages may also exist 
between guard and low or guard and chassis. As the largest 
voltage change allowed is between low and chassis, only 
shielding (no guarding) is incorporated on output lines; the 
resulting decrease in CMR is tolerable (the instrument's sys- 
tem specification, regardless of option or plug-in, is 80 dB 
at 60 Hz with a 1 k imbalance in either input lead). Guarding 
of the input lines is tolerable because of the reduced voltage 
allowed between low and guard. 

L, and L : (100 /<H and 220 ^H, respectively) provide a cur- 
rent step-up of approximately 2:1 — low to chassis. A high 
current, low-voltage pulse, on transfer, is forced into the 
transistor's base causing saturation and a resultant change 
in state of the latch used to provide storage. Programming 
transfer is accomplished in an identical manner — the 
grounds are simply reversed. The coefficient of coupling, 
M, is 0.3 to 0.5. 

It is imperative that the programming and digital output 
circuitry remain insensitive to externally changing voltage; 
otherwise, false triggering or programming, or a change in 
the output data could invalidate the measurement being 
made. As the low and earth grounds can vary by as much 
as 700 V (the maximum voltage that can be tolerated from 
low to guard is 200 V; 500 V from guard to chassis), and as 

A multilayer flex circuit is used to implement the shielding 
and guarding between the coils. The shields are constructed 
in a manner that eliminates any eddy currents that could be 
induced in a solid sheet that would result in unacceptable 
coupling losses. 

Double shields at different potentials are offset to reduce 


l Copr. 1949-1998 Hewlett-Packard Co. 


HP Model 3480A/B 
(With 3481 A Buffer Amplifier) 




90 days (25°C, <95% R.H.): 

±(0.01% of reading +0.01% of range) 

0"C to 55"C: ±(0.001% of reading +0.0003% ol range) per °C 

Reading Period: 950 /is. 

Reading Rale: Variable from 1 to 25 per s plus manual with front 

panel controls: 0 to 1000 per s with external trigger. 
RESPONSE TIME: 1 ms. Reads to'wlthln 1 count of final reading 
when triggered coincident with step Input voltage. 

COMMON MODE REJECTION: >80 dB, dc to 60 Hz with 1 kl} In 
either lead. 

HP Model 3480A/B 
(With 3482A DC Range Unit) 



±1000.0 mV. 
±10.000 V 
±100.0 V 
±1000.0 V 

OVERRANGE: 50% on all ranges. ±1200 V max input. 

RANGE SELECTION: Manual, automatic or remote. 

AUTOMATIC RANGING: Upranges at 140% of range: downranges at 

10% of range. 


Reading Period: 950 /:s 

Reading Rate (without range change): Variable from 1 to 25 per s 

plus manual with front panel controls: 0 to 1000 per s with ex- 
ternal trigger. 
Autorange Time: 

Filter Out: 4 ms per range change. 

Filter A: 200 ms per range change. 

Filter B: 1 s per range change. 
Response Time (without range change): 

Filter Out: 1 ms. Reads to within 1 count of final reading when trig- 
gered coincident with step Input voltage. 

Filter A: 200 ms to within 1 count of final reading. 

Filter B: 1 s to within 1 count of final reading. 
100 mV. 1000 mV, 10 V ranges: >10'°B. 
100 V, 1000 V ranges: 10M!) ±0.1%. 
of the peak common-mode voltage to the resultant error tn reading 
with kL' unbalance in either lead. 
DC: >80 dB. 
AC. (50-60 Hz): 

Filter Out: >80 dB. 

Filter A: >110 dB. 

Filler B: >160 dB. 
NORMAL MODE REJECTION (NMR)- NMR is the ratio of the peak 
normal mode signal to the resultant error in reading. 

Filter Out: 0 dB. 

Filter A: <30 dB at 60 Hz and above. 
Filter B: <80 dB at 60 Hz and above. 
Manual or Remote. 

HP Model 3480A/B 
(With 3484A Multifunction Unit) 



±1000.0 mV. 
±10.000 V. 
±100.0 V. 
±1000.0 V. 

OVERRANGE: 50% on all ranges. ±1200 V max input. 
RANGE SELECTION: Manual, automatic or remote. 

AUTOMATIC RANGING: Upranges at 140% of range; downranges at 
10% of range. 

Reading Period: 950 /is. 

Reading Rate (without range change): Variable from 1 to 25 per s 

plus manual with front panel controls: 0 to 1000 per s with ex- 
ternal trigger. 
Autorange Time: 

Filter Out: 4 ms per range change. 

Filter A: 200 ms per range change. 

Filter B: 1 s per range change. 
Response Time: (without range change): 

Filter Out: 1 ms to within 1 count of final reading when triggered 
coincident with step input voltage. 

Filter A: 200 ms to within 1 count of linal reading. 

Filter B: 1 s to within 1 count of final reading. 
100 mV, 1000 mV. 10 V ranges: >10'»Q. 
100 V, 1000 V ranges: 10M!i±0.1%. 
of the peak common-mode voltage to the resullant error In reading 
with 1 kG unbalance in either lead. 
DC: >80 dB. 
AC (50-SO Hz): 

Filter Out: >80 dB. 

Filter A: > 110 dB. 

Filler B: >160 dB. 
NORMAL MODE REJECTION (NMR): NMR is the ratio of the peak 
normal mode signal to the resullant error in reading. 

Filter Out: 0 dB. 

Filter A: >30 dB at 50 Hz and above. 
Filter B: >80 dB at 50 Hz and above. 
Manual or Remote. 

OHMS. Option 042 


1000.0 Q 
10.000 kS! 
100.00 ki> 

1000.0 fea 

10.000 M!7 
OVERRANGE: 50% on all ranges. 
RANGE SELECTION: Manual, automatic or remote. 
AUTOMATIC RANGING: Upranges at 140% of range: downranges at 
10% of range. 

90 days (25*C =5'C. <95% R.H.): 

1000'..' thru 1000 k!! ranges: ±(0.01% of reading +0.01% of range). 
100 O range: ±(0.02% of reading ±0 .05% of range). 
10 MO range: ±(0.1% of reading +0.01% of range). 
Reading Period: 950 /is. 

Reading Rate (without range change): Variable from 1 to 25 per 5 
plus manual with front panel controls; 0 to 1000 per s with ex- 
ternal trigger. 


© Copr. 1949-1998 Hewlett-Packard Co. 

Response Time (full scale step Input): 

100 I! thru 100 M! ranges (no tillering): 1 ms. Reads to within 1 

count ot tine! reading. 
100 kli range (Filter A): 200 ms to within 1 count ot final reading 
10 Mti range (Filter A): 2 s to within 1 count ot final reading. 
Note: Due to noise generated In the unknown resistance, filtering may 
be required tor quiet readings with Inputs > 100 k*.J. Response times 
with filtering are proportional less than those shown tor inputs below 
full scale. 

VOLTAGE ACROSS UNKNOWN: 1 V at full scale on all ranges. 
True rms ac Voltage Option 043 



1000.00 mV. 
10.000 V. 
100.00 V. 
1000.0 V. 

OVERRANGE: 50% on all ranges. 1500 V peak max input. 

RANGE SELECTION: Manual, automatic or remote. 

AUTOMATIC RANGING: Upranges at 140% of range: downranges at 

10% of range. 


Reading Period: 950 us. 

Reading Rate (without range change): Variable from 1 to 25 per s 
plus manual with front panel controls; 0 to 1000 per s with ex- 
ternal trigger. 

Response Time (full scale step Input, without range change): 
AC Coupled: 1 s to within 5 counts of final reading. 
DC Coupled: 15 s to within 5 counts of final reading. 


CREST FACTOR: 7:1 al full scale. 70:1 at 10% of full scale. 


Mainframes. Plug-ins and Options 
□C Ratio 3480A/B Option 002 
DISPLAYED RATIO: Display In all functions Is proportional to the ratio 
of the Input voltage to the external 10 V dc reference voltage ap- 
plied to rear-panel Ratio terminals. 
ACCURACY (with respect to external reference voltage): 
10 V or 100 V =5% external reference: Same as basic Instrument 

accuracy specifications. 
10 V, 100 V +5% lo +35% or 10 V. 100 V -5% to -13%: 
Add —0.02% ot reading to basic Instrument accuracy specifications. 
INPUT CHARACTERISTICS (ratio reference terminals): 

INPUT VOLTAGE: +10 V or +100 V (referenced to Low side of 

10 V Ratio Range: 100 kO ±1.5%. 
100 V Ratio Range: 100 kO ±0.5%. 


Remote controls are selected by application of a 'Low' state (logical 
'0') to the remote lines through a rear-panel connector. 

ENCODE (external trigger): Initiates a measurement period. Actu- 
ated by application of 'Low' state for >50 lis. Line must be in 
'High' state >50 us before applying 'Low' slate. Minimum time 
between ENCODE commands: 1 ms. 

INHIBIT (Interface Hold): Disables front-panel Sample Rate control. 

RATIO SELECT (Non-lsolaled Remote Control only): Selects Ratio 
Measurement (if mainframe has the Ratio option). 

FILTER SELECT (3482A. 3483A only): Selecls Filter A or Filter B. 
one line per filter. 

RANGE SELECT (3482A. 3484A only): Selects measurement range; 
one line per range. 

FUNCTION SELECT (3484A only)- Selects measuremeni function: one 
line per function. 

PROGRAM (3482A Option 021. 3484A Option 041 only): 
Accepts program commands when 'Low' state is applied for >50 us. 
Prevents changes in previously selected program when 'High' state 
is applied lor >50 us. Does not affect operation of ENCODE line 
A minimum of 1 ms must be allowed between PROGRAM and 
ENCODE commands. 

FLAG (Print Command): Line remains 'High' during reading period. 
Line changes lo 'Low' to Indicate completion of reading period 
and remains 'Low' until start of next reading period. 

PROGRAM FLAG (3482A Option 021. 3484A Option 041 only): Line 
remains 'Low' until Program Is executed. Line then goes 'High' 
upon execution, then 'Low' after programming Is completed (~1 ms). 

Non-Isolated Remote Control is standard on the 34S1A. 3482A and 

ISOLATED REMOTE CONTROL (3482A Option 021. 3483A Option 041) 
34B2A Option 021. 3484A Option 041 will operate only with 3480A/B 
mainframes equipped with Isolated Digital Output (Option 004). Iso- 
lated Remote Control for the 3481A is provided in the mainframe 
Isolated Digital Output option. 


The Digital Output Options provide measurement data outputs In digital 
form for printer and systems applications. In addition, input lines are 
included lo remotely control triggering of the 3480A/B. 
Non-isolated Digital Output Is available both as a factory-Installed 

option (3480A/B Option 003) and a field Installable accessory 

(HP 11147A). 

POWER: 115 V or 230 V ±10%, 50 Hz lo 400 Hz, 60 W max (including 
plug-in, options, normal environmental conditions). 

INPUT TERMINALS: High, Low and Guard terminals on both from 
and rear panels of 3481A, 3482A and 3484A. Front/Rear selector 
switch on fronl-panel of plug-In. High and Low Ratio Reference Input 
terminals on 3480A/B rear panel. Low Ratio and Low Inpul termi- 
nals are electrically common. 


34B0A Basic Instrument: 11 lbs. 12 oz (5,25 kg) 
Including Options: 12 lbs. 8 oz. (5,7 kg) 
Shipping: 17 lbs. (7.75 kg) 
3480B Basic Instrument: 12 lbs, 12 oz (5,71 kg) 
Including Oplions: 13 lbs. 8 oz. (6,15 kg) 
Shipping: 18 lbs, (8.1 kg) 
3481A Nel Weight: 2 lbs. 11 oz (1.2 kg) 

Shipping: 5 lbs. (2,3 kg) 
3482A Basic Instrument: 4 lbs. (1,8 kg) 
Including Oplions: 4 lbs, 4 oz, (1.9 kg) 
Shipping: 7 lbs. (3,15 kg) 
3484A Basic Instrument: 4 lbs, 6 oz (1.97 kg) 
Including All Oplions: 6 lbs, 2 oz. (2,76 kg) 
Shipping: 8 lbs, (3,6 kg). 

HP 11148A Plug-in Extender Cable for servicing all plug-ins. . S 45 

HP 11149A Remote Conlrol Cable for all plug-ins S 25 

The following accessories add oplional capabilities not Included 
with the basic Instrument. Optional capabilities which are not 
listed as accessories can be ordered only al the time of initial 
purchase. The Isolated Remote Accessory, HP 11151A. can be 
used only when the 34B0A/B has the Isolated Digital Output 
Option 004, which is no! available as an accessory. 

HP11147A Non-isolated Digital Output for 3480A/B S200 

HP 11151A Isolated Remote Conlrol for 3482A 

3484A (requires 3480A/B Option 004) $200 

HP 11152A Ohms Converter for 3484A 1200 

HP 11153A AC Converter for 3464A S800 


HP3480A Vz Module Main Frame {800 

HP 3480B Full Rack Width Main Frame (900 

Main Frame Options: 

Option 002 DC Ratio $200 

Option 003 Digital Output $200 

Option 004 Isolated Digital Output $375 

HP 3481A Buffer Amplifier (includes Single Range DC Voltage 
and Non-Isolated Remote Control).., $350 

HP 3482A DC Range Unit (includes 5 Range DC Vollage and 

Non-isolated Remote Control) $700 

Oplion 021 Isolated Remote Control (requires Main Frame 
with Option 004, HP 11149A Remote Cable furnishedl . $200 

HP 3484A Multifunction Unit (Includes 5 Range DC Vollage and 

Non-isolated Remote Conlrol) $900 

Option 041 Isolated Remote Control (requires Main Frame 
with Option 004. HP 11149A Remote Cable furnishedl . $200 

Oplion 042 Ohms Converter $200 

Option 043 True RMS AC Converter $800 


Loveland. Colorado 80537 


© Copr. 1949-1998 Hewlett-Packard Co. 

H. Mac Juneau 

Mac received his BSEE from 
Swarthmore College in 1961 
and his MSEE and Ph.D. from 
the University of Minnesota in 
1965 and 1967 respectively. 
After graduation Mac came to 
Hewlett-Packard and worked on 
ac converters for DVM's, a job 
which has kept him occupied 
for three years. 

Outside working hours, Mac 
spends his time woodcarving 
and welding. 



from Colorado 
the HP Honors 
Tau Beta Pi. 

Lee Thompson 

Lee Thompson received his 
BSEE degree from the University 
of Texas at Austin in 1966. 
He joined HP's Loveland Division 
that same year as a product 
designer. Lee did the product 
design and some circuit design 
on the rms converter In the 
HP 3450A before becoming 
involved with the true rms 
converter in the HP 3484A. where 
he has worked primarily on the 
amplifier design. 
* Lee received his MSEE degree 
State University in 1968 as a participant in 
Cooperative Program. He is a member of 


Craig Walter 

Craig earned a BSME from 
Stanford University in 1961, 
then continued at graduate 
school while working part time 
for Hewlett-Packard. After 
receiving his MSEE in 1963, 
Craig joined the HP Loveland 
Division. In 1968 he assumed 
responsibility for the four-digit 
DVM program. 

Craig's interests include 
sailing, bridge and wood- 
working. He is also an ardent 
sports enthusiast. 

HEWLETT-PACKARD JOURNAL ' January 1971 volume 22 . Number 5 

Hewlett-Packard S A 121 7 Meyrin - Gone**, Switzerland • Yohagawa- Hewlett-Packard Ltd . Shlbuya-Ku. Tokyo tSt Japan 

Editor R H Snyder Editorm Board R. P Oolan H L Robert* L D Sheroali* An Director Arvid A Oameraon A*»t»1*ni MandelJo'dan 

© Copr. 1949-1998 Hewlett-Packard Co.