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Issue 1/2008 
www.edn.com 




2007: a less-than- 
memorable year for 
DRAM Pg 74 

EDN.comment: Oper^- 
cellular-r^etwork boasts 
lack substar^ce Pg 10 

Signal Integrity: 

Initial cor^ditior^ Pg 22 

Design Ideas Pg 65 
Scope Pg 80 



COMMUNICATIONS- 
CENTRIC TEST GEAR 



I 



SYJVJBO 




Page 36 

RFI: KEEPING 
NOISE OUT OF 
YOUR DESIGNS 

Page 25 

NOVEL 
MEASUREMENT 
CIRCUIT EASES 
BATTERY-STACK- 
CELL DESIGN 

Page 47 



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Communications- 
centric test gear 
sharpens symbol 
recognition 

Designers pursue 
next-generation wire- 
V-y less developments 
with modulation-aware test tools, 
though evolving standards pres- 
ent problems from the PHY to the 
data layers. by Maury Wright, 
Editorial Director 




'III 

RFI: keeping noise 
out of your designs 

1^ Noise from cell 
y phones, digital oscilla- 
Victors, and even fluores- 
cent lights is assailing your elec- 
tronic designs. Learn what causes 
this noise and what you can do to 
increase your system's immunity 
to radio-frequency interference. 
by Paul Rako, Technical Editor 



DESIG 



IDEAS 



EDN 

contents 



110.08 




±L SINGLE-CELL 

BATTERY 



Novel measurement 
circuit eases battery- 
stack-cell design 

A A transformer and 
/I / diode on each cell 

I I allows isolated 
measurement, by Jim Williams and 

Mark Thoren, Linear Technology 



Use the MCLR pin as an output with PIC microcontrollers 
High-speed clamp functions as pulse-forming circuit 
Depletion-mode MOSFET kick-starts power supply 
Simple continuity tester fits into shirt pocket 
White LED shines from piezoelectric-oscillator supply 




JANUARY la 2008 | EDN 5 



Energy-Efficient Power 

Solutions 

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ORIng FET Power Rail Controller 


Replaces low-efficiency diodes with high-efficiency high-reliability 
control and protection solutions 


UCC28060 


Industry's First Single-Chip 
Interleaved RFC Controller 


Dual phase for high-efficiency high-power density and easy phase 
management for light-load efficiency 


UCD9112 


Digital Controller w/Configurable GUI 


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for servers, wireless infrastructure, datacom and telecom equipment 



TMS320F28335 Digital Signal Controller Highly integrated digital controller improves efficiency of renewable energy 

svstems 




contents no 



pulse 




Dilbert 16 



15 USB-bridge controller supports multilevel-cell 
NAN D flash 



15 ICs address speedy Ethernet devices' need for 
EMI, ESD protection 



16 Altera CPLD targets portable-system applications 



18 National Instruments aims at high-volume 
applications 



18 Integrated dc/dc regulators improve efficiency 
in server, embedded-system applications 

20 Research Update: Silicon fatigue: not a myth; IBM 
millimeter- wave wireless technology inches toward 
commercialization; STMicroelectronics claims first 
45-nm CMOS-RF chips; Silicon nanocrystals show 
promise for solar cells 




DEPARTMENTS 
& COLUMNS 

10 EDN.comment: Open-cellular-network boasts lack 
substance 



PRODUCT 
ROUNDUP 



Discrete Semiconductors: Power, N-channel, 
and switching MOSFETs 



22 Signal Integrity: Initial condition 



74 Supply Chain: 2007: a less-than-memorable year 
for DRAM; Demand-driven downturn possible in 
2008; California vetoes ROHS-expansion bill 



80 Scope: APEC 2008, a computer-based combat 
autopilot, and the credit crisis 



EDN® (ISSN#00 12-75 1 5), (GST# 123397457) is published biweekly, 26 times per year, by Reed Business Information, 8878 Barrons Blvd, Highlands Ranch, CO 80 129-2345. Reed Business Information, a division of Reed 
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Properties Inc, used under license. A Reed Business Information Publication/Volume 53, Number 1 (Phnted in USA) 



JANUARY 10, 2008 | EDN 7 



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ONLINE ONLY 

Check out these Web-exclusive articles: 

High-side current-sense-circuit problem 

Marcello Salvatierra, an applications engi- 
neer witli National Semiconductor's ampli- 
fier group, has observed a rather serious 
problem with the classic high-side current- 
sense circuit. 

■ www.edn.com/0801 1 0toci 



AMD: fading fast, nearly fini? 

■ www.edn.com/article/CA6513024 

Footprints of the EDA industry 

■ www.edn.com/080110toc2 



EDA place-and-route start-up ATopTech 
out of the gates with Broadcom win 

www.edn.com/article/CA6510888 

Equipment "quits": 
Beware bad wall warts 

www.edn.com/0801 lOtocS 



TSMC to qualify 65-nm 
mixed-signal, RF tools 

■ www.edn.com/article/CA6512752 

SOC yield management: an emerging 
issue that could reshape the industry 

' www.edn.com/080110toc4 



ON Semi to acquire AMIS for $91 5M 
to create power-semi giant 

www.edn.com/article/CA6512742 

Can Nokia compete with Apple's 
iPhone on music and "lifestyle?" 

• www.edn.com/080110toc5 



IMEC details high-k metal-gate planar 
CMOS progress 

www.edn.com/article/CA651 1456 




READERS' CHOICE 

A selection of recent articles receiving 
high traffic on www.edn.com. 

EDM Hot 1 00 Products of 2007 

ED/Vs editors offer up their annual list of 
the year's 1 00 most significant ICs, com- 
ponents, buses, boards, EDA tools, power 
devices, test instruments, and more, 
^www.edn.com/hotl 00 

Robots on the march: robotics 
platforms and development tools 

^ www.edn.com/article/CA6505566 



n 



1 



Avnet's "foundry phenomenon" 

^ www.edn.com/article/CA651 1351 



Protecting interfaces from ESD 

www.edn.com/article/CA6505573 

Identity crisis in the gray market 

^www.edn.com/article/CA6509284 



2008 capital spending to shake IC-in- 
dustry foundation, IC Insights reports 

^ www.edn.com/article/CA6509480 



High-brightness LEDs usher 

in new applications and standards 

www.edn.com/article/CA651 21 50 



IBM uses light to build 
supercomputers on a chip 

www.edn.com/article/CA651 01 84 

Decompensating amplifiers improve 
performance 

www.edn.com/article/CA6505576 



IBM, partners claim 32-nm high-k/ 
metal-gate SRAM, SOI 

-»www.edn.com/article/CA651 0863 



NEW BLOGGER 

HIRE GROUND 

In this new blog, you will read about a real- 
life engineering-job search as experienced 
by our unemployed guest blogger. Bill 
Betts. We encourage readers to share their 
own experiences and job-hunting 
advice with other engineers facing career 
and job changes. 
www.edn.com/hireground 



FROM EDM's BLOGS 
Embedded-processor recycling: 
Far out idea ... too far out 

From Leibson's Law, 
by Steve Leibson 

^ ^ The five professors who 

J authored this proposal 
^^■^^7 appear to be unaware 
^^k^^A '^^^ decade of 

^Kjgftl SOC development, the 
^^^^^H way that consumer-elec- 
tronic products are designed in the 
21 st century the way complex ICs are 
tested and characterized, or the way 
designers use ASSPs in their product 
designs. 

^ www.edn.com/0801 10toc8 

The Uncanny Valley: Beowulf's 
dead-end alley 

From Brian 's Brain, by Brian Dipert 
Flesh-and-blood 
Hollywood actors prob- 
ably don't have to worry 
about job security ... yet. 
But the time of cyber- 
reckoning is looming on 
the horizon, and I suspect it'll be here 
sooner than some of you believe. 
^ www.edn.com/0801 10toc9 




JANUARY 10, 2008 | EDN 9 



E D N . C O M M E N T 




BY MAURY WRIGHT, EDITORIAL DIRECTOR 



Open-cellular-network 
boasts lack substance 

It was really quite comical watching Verizon Wireless and AT&T 
argue in the mainstream media about the "openness" of their cel- 
lular networks. The problem is that neither company's approach 
is really open, and neither company has addressed what we need 
to spur real innovations that can leverage cellular radios: network 
technology that enables embedded cellular radios that companies 
can deploy at a reasonable price. 

If you missed the openness skir- 
mish, here's a quick summary. First, 
Verizon issued a press release (http:// 
news.vzw.com/news/2007/1 l/pr2007- 
ll-27.html) stating that, by the end 
of 2008, the company would support 
devices on its network that Verizon 
doesn't sell. Immediately, AT&T re- 
sponded in a USA Today article, claim- 
ing that its network has always been 
open (www.usatoday.com/tech/wire- 
less/phones/2007-12-05-att_N.htm). 

AT&T, I presume, is partially cor- 
rect. If you have a valid AT&T SIM 
card, you can put it in almost any un- 
locked GSM (global-system-for- mo- 
bile-communications) phone, and 
it will work. There are exceptions in 
the smartphone area. For instance, if 
you want your Blackberry to work to 
its potential, then you should buy it 
from AT&T. And even AT&T admit- 
ted that Apple's iPhone would remain 
locked to the network. AT&T did say 
that it will unlock phones for custom- 
ers that either fulfill their contracts or 
pay full price for a phone. That ap- 
proach is new, but Internet hacks that 
unlock phones are widely available, 
and many local phone shops will un- 
lock a GSM phone for a fee. 

As for Verizon, details of its open 



There are dozens 
if not hundreds of 
products that might 
be more compelling 
if they integrated a 
cellular radio. 

plan were still to come at press time. 
But the press release notes that a third 
party that wants to offer a device for 
the Verizon network must have a new 
Verizon lab certify that device. I'd guess 
that a few such devices are already in 
that certification process, but wide 
choice probably will not happen soon. 
Verizon didn't say whether it would 
support CDMA (code-division/multi- 
ple-access) phones sold on competing 
CDMA networks. The CDMA com- 
munity has never relied on SIM cards, 
although ironically, most CDMA chip 
sets offer SIM-card support. 

But here's what is missing from both 
announcements: There are dozens if 
not hundreds of products that might 
be more compelling if they integrated 
a cellular radio. Take a handheld prod- 
uct such as the Sony PSP (PlayStation 
Portable). The PSP has Wi-Fi, but a 
cellular link would provide an "any- 



where" connection to support multi- 
player games or content downloading. 
You can make the same case for MP3 
players. What about GPSs? Dash Nav- 
igation has integrated a cellular radio 
in its GPSs so that autos can send real- 
time traffic data back to a database 
that serves all Dash owners. 

Today, a device with an integrat- 
ed cellular radio needs an account 
and phone number just like a phone. 
But most embedded cellular applica- 
tions don't need a phone number and 
would use only the data services that 
the mobile carrier supports. 

The real issue, I suspect, is price 
and not technology. My family has an 
AT&T Wireless account, and we pay 
$6 for each additional phone num- 
ber. That cost isn't much when we 
need only three phone numbers. But 
what if I had another dozen cellular ra- 
dios in a variety of portable electron- 
ics? There is no way I would pay $6 
for each. Nor would I buy an $80-per- 
month data contract specific to an em- 
bedded radio. 

Now, there are applications that will 
pay the going rate. A cellular-linked 
portable medical device that helps 
keep a consumer alive is clearly worth 
a dedicated account. But to spur inno- 
vation and an explosion of new mobile 
radios in everyday devices, the carri- 
ers need to develop a technology and 
a business plan that make support for 
those devices less costly. I'll buy one 
relatively expensive wireless-data ac- 
count, but the incremental cost to use 
that service on multiple devices needs 
to be almost free. And this approach 
would benefit the carriers. Data servic- 
es aren't selling as well as they'd like. 
Meanwhile, the carriers pursue newer, 
faster data technologies.EDN 

Contact me at mgwright@ednxom. 



ATEDN.COM 



\±} Go to www.edn.com/0801 1 0ed 
and click on Feedback Loop to post 
a comment on this column. 



10 EDN I JANUARY 10, 2008 



SEMICONDUCTORS 



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Rarely Asked Questions 



Half Full or Half Empty? Thoughts on Capacity. 

Q. Are all components with just two wires as 

connplicated as the resistors we discussed recently? 



A. Capacitors certainly are. 

If you put a pint of liquid into a quart pot, 
the optimist will declare it half full, but on 
the other hand the pessimist will complain 
that it's half empty. 

Engineers, on the gripping hand', know that 
the glass is too large. 

It's a matter of capacity. Capacitors, like the 
resistors we discussed recently, are more 
complex than the simplicity of their two leads 
suggests, and bigger is not necessarily better. 

A capacitor has more characteristics than its 
capacity and maximum operating voltage. 
In parallel with the nominal capacity there 
will be leakage resistance and dielectric 
absorption. In series, there will be induc- 
tance and effective series resistance (ESR). 
ESR is important; the largest unintentional'' 
explosion I ever caused occurred when I 
was working on ultrasound cleaners and 
replaced a faulty mica high-frequency (HP) 
capacitor in the tank circuit of a 5-kW ultra- 
sonic generator with a high-ESR oil-filled 
capacitor. I was lucky to survive, but the 
ultrasonic generator didn't. 

Even low-frequency integrated circuits 
(ICs) contain transistors with a frequency 
response of hundreds or thousands of MHz. 
If the supply pins of the IC are not short- 
circuited at HP, the parasitic components 
formed by the printed circuit board tracks 
may create resonators and oscillate — pos- 
sibly at such a high frequency that the oscil- 
lation is not visible on an oscilloscope. The 
capacitor used to create this HP short-cir- 
cuit must have low inductance — and short 
leads. Too large a capacitance is unlikely to 
have sufficiently low inductance, and some 
types of small capacitors (such as spiral- 
wound plastic film) may still be unsuitable. 
On the other hand the precision and stabil- 
ity of such decoupling capacitors are com- 
paratively unimportant. 




In active filters, precision and stability are 
of overriding importance. In low-frequency 
supply decoupling, the ability to handle 
high ripple currents without overheating 
limits the types we may use. 

Pifteen or twenty years ago the dielectric 
absorption of capacitors for use in sample 
and hold (SHA, S/H or, sometimes, T/H 
[track and hold]) circuits was very important. 
It is still important for SHAs that use dis- 
crete capacitors, but today these capacitors 
are usually integrated onto a chip, and are 
not separate components. Leakage is still 
important in RC timing circuits, though. 

Even the difference between polarized 
and unpolarized capacitors is important 
in ac applications. 

Choosing capacitors involves a lot more than 
simply calculating the required capacitance. 
The linked article discusses the issues in 
much more detail. 

i To learn about the Motie race, and their additional 
gripping hand, read "The Mote in God's Eye" (ISBN 0- 
671-21833-6) and its sequel "The Gripping Hand" (ISBN 
0-671-79573-2) by Larry Niven and Jerry Pournelle or see 
h ttp -.//en.wikipedia.o rg/wi ki/G ripping_hand 

ii Don't ask about my experiences with pyrotechnics and 
blasting gelatine. 



To learn more 
about Capacitors 

Go to: http://rbi.ims.ca/5697-101 



Contributing Writer 
James Bryant has been 
a European Applica- 
tions Manager with 
Analog Devices since 
1982. He holds a degree 
in Physics and Philoso- 
phy from the Univer- 
sity of Leeds. He is also 
C.Eng., Eur.Eng., MIEE, 
and an FBIS. In addi- 
tion to his passion for 
engineering, James is 
a radio ham and holds 
the call sign G4CLF. 



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Design Engineer, Cisco 

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EDITED BY FRAN GRANVILLE 




INNOVATIONS & INNOVATORS 



USB-bridge controller supports 
multilevel-cell NAND flash 



Last year, Cypress Semiconductor 
announced the West Bridge fam- 
ily of chips that primarily add high- 
speed-USB support to products such as 
mobile handsets. Full-speed transfers be- 
tween handset memory and the USB inter- 
face, with no assistance from the handset 
application processor, have proved to be 
the primary benefit of West Bridge. Now, 
Cypress is launching the Astoria flavor of 
West Bridge, adding an MLC (multilevel- 
cell)-NAN D-f lash-memory controller. 

Cypress claims that MLC flash costs a 
third of single-level-cell flash for the same 
storage capacity, making MLC support at- 
tractive in applications ranging from hand- 
sets to media players to digital cameras. 
Many of the standard storage modules. 



such as USB-memory sticks or SD (Se- 
cure Digital) cards, use MLC memory but 
integrate the controller, thereby hiding the 
complexity of the control function. But de- 
signers who wish to embed MLC flash 
have little choice when it comes to the 
control function. 

Astoria can control 16 MLC NAND 
memories and supports devices from all 
of the major flash vendors. The controller 
includes bad-block management, static- 
wear-leveling support, and 4-bit ECC (er- 
ror-correction code). You can interface As- 
toria to all popular processors and DSPs. 
And, like its predecessor, the chip supports 
1 6 USB endpoints and a host of program- 
mable-l/0 features. The IC sells for less 
than $5 (500,000).-by Maury Wright 




The Cypress West Bridge Astoria chip 
provides a three-way interface among 
a processor, USB, and memory and includes 
a multilevel-cell-NAND-flash controller. 

Cypress Semiconductor, www. 
cypress.com. 



ICs address speedy Ethernet devices' need for EMI, ESD 



The Ethernet communica- 
tion standard is morphing 
into 10/100/1000-Mbps 
versions and gaining inher- 
ent power-delivery capability 
in POE (power-over-Ether- 
net) designs. System design- 
ers are responding with fea- 
ture-packed products, such 
as VOIP (voice-over-Internet 
Protocol) phones and IP cam- 
eras. However, it is becoming 
increasingly difficult to protect 
these devices from EMI (elec- 
tromagnetic-interference) 
noise and ESD (electrostatic 
discharge) without degrading 




The AS1 601/02 EMI/ESD- 
suppression ICs actively 
protect Ethernet devices 
from noise and damaging 
bursts of electrical energy. 

system performance. Passive 
devices, including TVS (tran- 
sient-voltage-suppression) 
diodes for ESD protection 



and external chokes for EMI, 
offer some protection but 
often at the expense of sys- 
tem performance. Address- 
ing these problems, Akros 
Silicon has introduced two 
devices that it claims are the 
first active devices for sup- 
pressing EMI and ESD and 
promoting EMC (electromag- 
netic compatibility). 

The integrated AS1602 
common-mode-EMI-noise- 
and ESD-suppression IC pro- 
vides less-than-0.5n com- 
mon-mode impedance to 
ground and complies with Eth- 



protection 

ernet-performance specs. The 
CMOS chip goes between the 
Ethernet PHY (physical) layer 
and the line transformer. Exter- 
nal ferrites steer bursts of en- 
ergy to the chip, which quickly 
sinks the transient energy; in 
less than 1 nsec, diodes steer 
the energy away from the PHY 
layer.The AS1 601 version pro- 
vides only EMI suppression. 
The AS1602 and AS1601 
CMOS ICs sell for $1.60 and 
$1.23 (1000), respectively 

-by Margery Conner 
Akros Silicon, www. 
akrossilicon.com. 



JANUARY 10, 2008 | EDN 15 



)Lise 



Altera CPLD targets 
portable-system applications 



With an eye on win- 
ning more sockets 
in low-power-por- 
table-system applications and 
a greater share in the FPGA 
market, Altera has announced 
the Max IIZ, the latest member 
of its low-power Max II non- 
volatile-CPLD lineup. Altera 
based the "zero-power" Max 
IIZ on the Max II four-look-up- 
table architecture, which the 
company implemented in the 
TSMC (Taiwan Semiconductor 
Manufacturing Co, www.tsmc. 
com) 1 .8V, 0.1 8-nm flash tech- 
nology Altera will initially offer 
the device in the 240-logic- 
element EPM240Z and the 
570-element EPM570Z with 



I/O counts of 54 to 260. 

One of the key selling points 
of the device is its low power, 
according to Dennis Steele, di- 
rector of product marketing for 
the Max II family. The device 
boasts typical standby power 
of 29 [xA and maximum power 
of 1 50 [jlA. To achieve these 
numbers. Altera tweaked the 
architecture's power-on-re- 
set circuitry to reduce stand- 
by-power consumption and 
slightly increased the device's 
transistor-threshold voltage to 
reduce standby leakage. 

Traditionally portable-system 
designers have only sparing- 
ly used low-power CPLDs as 
glue logic because program- 



mable logic tends to be pow- 
er-hungry Designers typically 
use CPLDs to handle com- 
munications and, sometimes, 
power management between 
a system's main processor and 
other devices. The Max IIZ, 
however, boasts a combina- 
tion of low power, reasonably 
high logic-element counts, ad- 
vanced performance, and high 
I/O counts, allowing designers 
to use low-cost CPLDs for a 
broader number of essential 
tasks in portable-system ap- 
plications. For example, the de- 
vice can perform as a voltage- 
level shifter, expand general- 
purpose I/O, act as a bridge, 
serve the processor with extra 



VOLTAGE- 
LEVEL SHIFTER 








GENERAL- 
PURPOSE-I/O 
EXPANDER 








BRIDGE 








MAX IIZ FUNCTIONS 



The Max IIZ can perform as a voltage-level shifter, expand general-purpose I/O, act as a bridge, 
serve the processor with extra I/O, and function as an interface between the processor and the 
system peripherals. 



DILBERT By Scott Adams 





HOU WAS 




THE ALL 




HANDS 


CREEPY. 


rAEETING? 





a Altera 
tweaked 
the architec- 
ture's power- 
on-reset circuitr/ 
to reduce 
standby-power 
consumption. 

I/O, and function as an inter- 
face between the processor 
and the system peripherals. 
The device also has an oscil- 
lator, which other CPLDs lack, 
according to Steele. 

This mix of functions allows 
designers to configure a Max 
IIZ to work as a low-power co- 
processor in handheld devices 
that mix, for example, phone 
and MPS functions. One of the 
biggest problems handheld-de- 
vice designers face is that, al- 
though a device may have long 
battery life when a consumer 
uses it solely as a phone, it 
quickly sucks up battery pow- 
er when the user employs it to 
play music. This consumption 
typically occurs because the 
ARM (www.arm.com)-based 
SOC (system on chip) that con- 
trols the system must run at full 
speed to communicate with the 
audio codec and media FIFO. 
The Max IIZ's internal oscillator 
allows designers to configure 
the CPLD as a coprocessor 
that includes and controls the 
codec and media-FIFO func- 
tions at lower power than the 
ARM-based SOC, thus sav- 
ing overall battery power. The 
company offers each Max IIZ 
device in 5x5-, 6x6-, 7x7-, 
and 1 1 XI 1-mm MBGA pack- 
ages. The EPM240Z M68 will 
cost $1.25 (1 million), and Al- 
tera will begin shipping it this 
quarter. 

-by Michael Santarini 
[> Altera Corp, www.altera. 
com. 



16 EDN I JANUARY 10, 2008 



uoich 



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)Lise 



National Instruments aims at high-volume applications 



Integrating hardware that 
includes an embedded real- 
time processor and a re- 
configurable FPGA, National 
Instruments' new cRIO-9072 
and cRIO-9074 CompactRIO 
systems target high-volume 
industrial applications. The 
systems extend NTs FPGA- 
based deployment platforms 
that share common hardware 
architecture and I/O modules. 
Using this standard architec- 
ture and LabView FPGA and 
Real-Time tools, engineers can 
design and prototype industri- 
al-monitoring-and-control ma- 
chines with PXI (peripheral- 
component-interconnect-ex- 
tensions-for-instrumentation), 
PC, or standard CompactRIO 
hardware and then move to 



the cRIO-907x CompactRIO 
systems to reduce deployment 
costs. Because engineers can 
reuse the same LabView code 
during prototyping and deploy- 
ment, they can shorten time to 
market and increase machine 
reliability 

To reduce costs, the cRIO- 
907x integrates the proces- 



sor and the FPGA chip on 
the same PCB (printed-circuit 
board). The cRIO-9072 system 
combines an industrial, 266- 
MHz real-time processor; 64 
Mbytes of DRAM; 1 28 Mbytes 
of nonvolatile storage; and an 
eight-slot chassis with a recon- 
figurable, 1 million-gate FPGA 
chip. Prices start at $ 1 999. The 



cRIO-9074 contains a 400- 
MHz real-time processor, 128 
Mbytes of DRAM, 256 Mbytes 
of nonvolatile storage, and an 
eight-slot chassis with a 3 mil- 
lion-gate FPGA chip. Prices 
start at $2999. 

—by Warren Webb 
National Instruments, 
www.ni.com/compactrio. 




The new cRIO-907x CompactRIO systems integrate a controller and a chassis; the integration 
lowers recurring costs for high-volume OEM applications. 




18 EDN I JANUARY 10, 2008 





More 16-bit DAC performance for more designs. 
In process control, analog is everywhere. 



t1LSB IMLr 



LDAC and CLR 
Pin Functionality 



Small Packages: 
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Power-On Reset to 
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AD5064 

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The first low voltage quad with ±1 LSB INL @ 16 bits. 
Unmatched accuracy and pin functionality, combined. 



ISOLATION 



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Open-Loop and Closed- Loop Applications 

Our newest portfolio of industrial 16-bit DACs offers best-in-class 
performance, and a world of system configuration possibilities. You'll find 
single, dual, and quad option DACs, in small packages, with a full range of 
design tools and support. In addition to these DACs, Analog Devices offers 
hundreds of other IC solutions to meet all your process control needs. For 
more information, visit www.analog.com/16-bitDACs or call 1-800-AnalogD. 



AD5754 

Flexible Solution for Closed- Loop Systems 

The AD5754 provides a software selectable output range of 5 V, 
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I— — < 1 ADC^ ^ ^MUXj^-^MPl^ 



Part No. ■ 




Price 


Ideally Suited to Open-Loop 


AD5060 


Single, 5 V,±1 LSB INL (max), 1 mA@5V 


$7.50 


AD5065 


Dual,5V, ±1 LSB INL (max), 2.3 mA@5V 


$11.25 


AD5064 


Quad, 5 V, ±1 LSB INL (max), 5 mA @ 5 V 


$15.95 


AD5764 


Quad, ±15V, ±1 LSB INL (max) 


$35.70 


Ideally to ^lilllllHIIIB flillliillillllillllll 


AD5752 


Dual, software-programmable output range of 5 V, 1 V, 
±5V, ±10Vin 24-leadTSSOP 


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AD5754 


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±5V, ±10Vin24-leadTSS0P 


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AD5664R 


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$10.45 



All prices shown are $U.S. at 1k quantities unless ottierwise noted. 
Aii parts 16-bit resolution. 




analog Is everywhere: 



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ANALOG 
DEVICES 



pulse 



RESEARCH UPDATE 



BY MATTHEW MILLER 



Silicon fatigue: not a myth 



In work with ramifications 
for MEMS (microelectro- 
mechanical systems), re- 
searchers at NIST (National 
Institute of Standards and 
Technology) claim to have 
proved that, contrary to con- 
ventional wisdom, bulk silicon 
crystals are vulnerable to fa- 
tigue from cyclic stresses. 

The scientists used 3-mm- 
diameter tungsten-carbide 
spheres to apply pressure to 
the surfaces of test crystals. 
Simply pressing, even for days 
at a time, caused no discern- 
ible damage. But cycling the 
test hundreds of thousands 



of times, even at low pressure, 
resulted in a gradually wors- 
ening damage pattern. 

NIST claims that this clear 
evidence of mechanical stress 
resolves a debate about 
cracks, which scientists ob- 
served in some MEMS struc- 
tures, by ruling out a com- 
petitive theory that fingered 
chemical corrosion as the cul- 
prit. The team proposes that 
its test found damage that 
conventional tensile-strength 
tests miss because it induced 
shear stress-causing crystal 
planes to slide against each 
other. And shear stress, the 




A silicon crystal exhibits increasingly obvious mechanical damage 
after 1 000 (a), 5000 (b), 20,000 (c), and 85,000 (d) cycles of a 
stress test. 



team notes with concern, is 
not uncommon in real-world 
applications. The next step for 
the research team is to scale 
down the testing from the cur- 



rent scale of hundreds of mi- 
crons to the submicron level. 

National Institute of 
Standards and Technol- 
ogy, www.nist.gov. 



IBM mil 11 meter- wave wireless technology 
inches toward commercialization 



IBM and Taiwan-based 
fabless chip maker MediaTek 
recently announced an 
agreement to develop chip 
sets based on IBM Labs' 60- 
GHz mmWave wireless tech- 
nology. Prototype mm Wave 
chips that IBM unveiled 
in early 2006 achieved 
throughput of 630 Mbps 
with a maximum range of 
1 0m, but the company touts 
the technology as potentially 
suitable for even higher 
bandwidth applications, such 



Prototype mmWave chips that 
IBM unveiled in early 2006 
achieved throughput of 630 
Mbps with a maximum range 
of 10m. 




as ferrying uncompressed 
high-definition video streams 
from set-top boxes to dis- 
plays. In this application, the 
limited range, which arises 
due to oxygen absorption, 
would be a feature, not a bug. 
Hollywood studios and other 
content owners prefer not to 
have their valuable content 
traveling in uncompressed 
form for long distances. 

In addition to chip develop- 
ment, IBM is researching 
IC packaging and antenna 
designs for the 60-GHz 
band. The IEEE (www.ieee. 
org) is working toward stan- 
dardization of applications in 
the 60-GHz band through 
the IEEE 802.1 5.3c working 
group. For more information, 
visit www.research.ibm. 
com/mmwave. 
>IBM Corp, www.ibm.com. 
>MediaTek, www.mtk.com. 
tw. 



STMicroelectronics claims 
first 45-nmCMOS-RF chips 

STMicroelectronics has announced the production of its first 
fully functional ICs using a 45-nm CMOS-RF process. The 
prototype devices integrate a complete signal chain, from 
detection of an RF signal through digital output, according 
to the company Measuring 0.45 mm square, including a low- 
noise amplifier, a mixer, an ADC, and filtering, the devices 
operate at 1.1V. 
STMicroelectronics, www.st.com. 



SILICON NANOCRYSTALSSHOW 
PROMISE FOR SOLAR CELLS 

Speaking of qualities of silicon that scientists once thought 
nonexistent (see "Silicon fatigue: not a myth," above), 
researchers at the US Department of Energy's National 
Renewable Energy Laboratory have shown that silicon 
nanocrystals exhibit multiple-exciton generation, mean- 
ing that they emit more than one electron per absorbed 
photon. In theory, solar cells harnessing this effect could 
achieve efficiency of 44% using sunlight and as much as 
68% using sunlight that lenses or mirrors have concen- 
trated. Conventional solar cells deliver 33 and 40% effi- 
ciency, respectively 

National Renewable Energy Laboratory, www.nrel. 
gov. 



20 EDN I JANUARY 10, 2008 



5 million reasons why SAR ADCs mean ADI. 
In motion control, analog is everywhere. 




Illlllll 




llllllll 




AD7356 
12-bit ADC 

• Simultaneous sampling 

• 5 MSPS per channel 

• Zero latency 

• 35 mW at 5 MSPS 

• 16-leadTSSOP package 

• 1k price: $7.89 

AD8138 
ADC Driver 

• Single-ended to differential conversion 

• Low distortion, -94 dBc SFDR @ 5 MHz 

• Fast settling to 0.1% in 16 ns 



At 5 MSPS and 35 mW power, this 
12-bit SAR ADC makes clioosing ADI easy 

ADC leadership is about optimizing more dimensions of converter 
performance and value for the unique requirements of each 
application. That is what our new AD7356 is all about. At 12 bits 
and 5 MSPS, it is the industry's fastest SAR ADC— by far. But it also 
offers zero latency, differential input, greater functional integration, 
simultaneous sampling, and just 35 mW power dissipation— all in 
a small 16-lead TSSOP footprint. 

The AD7356 eliminates the power consumption and latency concerns 
of flash and pipeline ADCs, and the size and cost drawbacks of 
multiple ADCs. So you get 5 MSPS throughput for one or two 
channels, while simplifying your design and lowering overall cost. 
The AD8138, with its low distortion and fast settling time, is a 
recommended ADC driver for maximum precision signal processing. 

For more information on the AD7356 and other leading SAR ADCs, 
please visit www.analog.com/SAR5MSPS or call 1-800-AnalogD. 



analog is everywhere: 



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□ ANALOG 
DEVICES 



SIGNAL INTEGRITY ■ 



BY HOWARD JOHNSON, PhD 



Initial condition 

The PCB (printed-circuit-board) transmission line in Figure 
la lacks an end termination. If you leave switch closed 
for a long time, the line comes to rest in a state with pre- 
cisely OV at all points. The transmission line in Figure lb 
behaves similarly. With closed, it also comes to rest in a 
state with OV at all points. The voltages in the two situa- 
tions are the same, but the currents differ. In the first case, the line at rest 
carries no current. In the second case, the end termination supplies a sub- 
stantial current as long as you hold the line in a low state. To make the 
numbers easy, assume 2V logic, a per- i current to OA. The combination of 



feet switch at the source, and values of 
lOOa for both and R^. Those val- 
ues produce a steady-state current of 
20 mA for Figure lb. 

At time zero, with both lines in 
their respective steady-state condi- 
tions, open both switches. In the case 
of Figure la, just before you open the 
switch, no current flows through it. 
Opening therefore changes noth- 
ing; it has no effect on the circuit. 
Opening in Figure lb has a dif- 
ferent effect. In the steady-state con- 
dition, 20 mA spills continuously 
through the switch. When you inter- 
rupt that state of events by opening 
S^, the current at the left end of the 
line changes from 20 mA to OA. You 
can emulate that effect with a super- 
position of two linear-current sources, 
Ig and l^, which connect (Figure Ic). 

Current source Ig replaces the 20 
mA of steady-state current flowing 
through in Figure lb. It sets the 
initial conditions before your switch- 
ing event, and it perpetually sinks 
20 mA. At time zero, a 20-mA step of 
current from source I^^ cancels the cur- 
rent from source Ig, bringing the net 



two sources duplicates the conditions 
at the left of Figure lb the moment 
opens. The linear-current-source 
model clarifies the actions that occur 
at time zero. Directing a positive step 
of 20 mA into the line must create a 
positive-step-voltage waveform mov- 
ing to the right with an amplitude of 
20 mAx50H=lV. In a 2V system, 
that scenario makes a half-sized step. 

If your goal is to inject a total volt- 
age step of 2V into the line, making 
a full-sized step, which initial state do 
you prefer? Starting with Figure la — 
that is, with no termination — you 
must do all the work with the top half 
of your totem-pole driver, sourcing a 
full 40 mA to create a full-sized signal. 
Most drivers can't source that much 
current. On the other hand, a circuit 
with a symmetric end termination en- 
joys the benefit of sinking 20 mA the 
entire time it holds low. When the 
bottom half of the totem-pole driver 
lets go, the line voltage at the source 
automatically jumps up halfway. The 
top half of the totem-pole driver then 
needs only to source the other half of 
the current (20 mA) to bring the line 



(a) 



1 



20 mA I 

(b) 




(c) 

Figure 1 End termination R/R^ estab- 
lishes an initial current before switching 
high. If you leave switch closed for 
a long time, the line comes to rest in 
a state with precisely OV at all points 
(a). The end termination supplies a 
substantial current as long as you hold 
the line in a low state (b). When you 
interrupt that state of events by opening 
the current at the left end of the line 
changes from 20 mA to OA. You can 
emulate that effect with a superposition 
of two linear-current sources, Ig and 1^, 
which connect (c). 



up to full voltage. A symmetric end 
termination biases the line at a half- 
way voltage, so that the driver need 
source or sink only enough current to 
swing the line halfway either direc- 
tion. That's why I like it.EDN 



MORE 



AT EDN.COM 



\±} Go to www.edn.com/0801 1 Ohj 
and click on Feedback Loop to post 
a comment on this coiumn. 



Howard Johnson, PhD, of Signal Con- 
sulting, frequently conducts technical 
workshops for digital engineers at Oxford 
University and other sites worldwide. 
Visit his Weh site at www.sigcon.com or 
e-mail him at howie03@sigcon.com. 



EDN I JANUARY 10, 2008 



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NOISE FROM CELL 
PHONES, DIGITAL 
OSCILLATORS, AND 
EVEN FLUORESCENT 
LIGHTS IS ASSAILING 
YOUR ELECTRONIC 
DESIGNS. LEARN WHAT 
CAUSES THIS NOISE 
AND WHAT YOU CAN 
DO TO INCREASE 
YOUR SYSTEM'S 
IMMUNITY TO RADIO 
FREQUENCY 
INTERFERENCE. 




OUT OF YOU 



BY PAUL RAKO • TECHNICAL EDITOR 

Steady streams of RF energy constantly engulf your 
electronic system. Some of this energy comes 
from the accidental byproduct of a system; oth- 
er RF sources, such as radios and radar, inten- 
tionally radiate energy. Some RF sources are so 
strong and so insidious that they create noise in 
simple wires, such as the magnet wire that forms 
the voice coil of a speaker. It is merely annoying 
for consumers to hear noise in their home-audio 
systems. However, RF noise that causes a machine to go haywire 
or an airplane's instruments to malfunction could imperil or even 
kill people. For this reason, the European Union and the Unit- 
ed States instituted RFI (radio-frequency-interference) testing for 
products that vendors sell there. When the European Union more 



than a decade ago instituted the CE 
(Conformite Europeenne) immunity- 
compliance tests, engineers soon learned 
that passing them is more difficult than 
passing the US FCC (Federal Commu- 
nications Commission) noise-radiation 
tests- "Engineers don't think it is a prob- 
lem until it is a problem for them," says 
Steve Bible, Microchip Technology's 
technical'Staff engineer. "They are in a 
real time crunch. They have made a bad 



design, and it's hard to convince them 
that it's bad. They want to find that one 
silver bullet — the one thing they can do 
so they can pass — except there is no sil- 
ver bullet." 

To provide your systems with robust 
RFI immunity, you must understand 
just how many RF sources your system is 
subject to. The electric-power industry 
broadcasts 50- or 60-Hz radio waves as 
it sends power to your house. Your watch 



JANUARY 10, 2008 | EDN 25 



has a 3 2 -kHz crystal that emits energy. 
Electronic ballasts for fluorescent lights 
operate at 40 kHz. Traffic lights use loop 
sensors that energize at 50 to 100 kHz. 
At higher frequencies, you soon en- 
counter "intentional radiators," which 
the FCC defines as radio stations; TV 
stations; and various private, public, and 
military radios, some of the most trou- 
blesome of which are cell phones. Ra- 
dar systems and exotic military systems 
lie even beyond cell phones on the fre- 
quency spectrum. Cosmic rays also cause 
problems (Reference 1). It is difficult to 
help a customer with an RFI-susceptibil- 
ity problem because hundreds of ways 
exist to hook up an amplifier in a sig- 
nal path, according to Steve Sockolov, 
product-line director for Analog De- 
vices' precision-linear-products group. 
You also must worry about a continu- 
um of source frequencies. To help cus- 
tomers with precision-measurement cir- 
cuits. Analog Devices has developed the 
AD8556 sensor-signal amplifier, a func- 
tional equivalent of the AD8555 ampli- 
fier, except that the AD8556 has 
EMI (electromagnetic-interfer- 
ence) filters on the input pins, 
the reference pin, and the clamp 
pin. These filters help suppress 
RFI across a wide range of fre- 
quencies. 

Not all RFI sources are causes 
for concern. The aforemen- 
tioned watch crystal operates at 
a relatively low frequency and 
transmits minuscule power lev- 
els. Other sources may or may 
not be problematic. For exam- 
ple, you may use a FET as a low- 
side switch in a synchronous 
buck regulator. The FET's pack- 
age case connects to the switch 
node and swings the entire pow- 
er-supply voltage (Figure 1). Be- 
cause this node operates at the 
power-supply frequency, you 
would think that it would radi- 
ate RF energy, but it may not. To 
radiate RF, current must be flow- 
ing. By using the package pin to 
carry the current and using the 
package tab to absorb the heat of 
the circuit, a clever designer can 
cool the FET and minimize RF 
radiation. 

One way to solve an immu- 
nity problem is to stop the RF 



AT A GLANCE 

□ RF sources can transmit energy 
into cables, PCB (printed-circuit- 
board) traces, and ICs. 



EI Symptoms of RF susceptibility 
can be tricky to diagnose. 



El The best technique is to kill noise 
at its source. 



EI Shielding is a high cost 
Band-Aid. 



El Careful layout and good system 
design can provide the best pro- 
tection from RFI (radio-frequency 
I interference). 

source. Automotive engineers decades 
ago learned this technique when they 
first installed radios into automobiles 
(see sidebar "Insidious RF" at the Web 
version of this article at www.edn.com/ 
OSOllOdf). It soon became evident that 
barring noise from the radio was a dif- 
ficult process, whereas killing the noise 
at its source was an effective technique. 
The engineers achieved this goal by 




HIGH CURRENTS 



Figure 1 A copper pour forms a large heat sink that may 
look problematical from an EMI perspective. Because it 
carries no current, however, the heat sink does not radiate 
large amounts of RF energy. 



adding capacitors to the alternator. The 
capacitors suppressed the diode-switch- 
ing spikes, minimizing circulating cur- 
rents and, thus, noise (Reference 2). 
The use of these techniques, along with 
tight layouts, will help you pass FCC ra- 
diation tests. Using these methods also 
subtracts one source of RFI that may 
cause immunity problems. 

The biggest problem in RFI arises be- 
cause you often have no control over the 
RF source that is polluting your system, 
such as the source you encounter in cell 
phones, which operate at high frequen- 
cies. This RFI can enter many parts of 
your design: the cables, the PCB (print- 
ed-circuit-board) traces, and even the 
ICs themselves. In addition, cell phones 
are everywhere, often sitting next to or 
atop your design while you are working 
on it. A few anecdotes tell the story: Bob 
Thomas, an engineer with Cisco Sys- 
tems, reports that, when he sets his cell 
phone in the package tray of his 2006 
Honda, the noise it radiates into the ra- 
dio is louder than the music that the ra- 
dio emits when it is on. Anoth- 
er Cisco engineer, Steve Abe, 
notes that placing his cell phone 
on his Palm Zire causes the Zire 
to reboot whenever he receives 
an incoming call. Francis Lau, 
an engineer with EM -transmitter 
manufacturer Aerielle, says that 
the stereo in his home makes a 
buzzing sound when he is about 
to get a call on his cell phone. 

To understand why cell phones 
can be sources of RFI at audio 
frequencies, you must examine 
the RF-transmission protocols. 
The NADC (North American 
digital-cellular) -phone system 
uses the TDM A (time-divi- 
sion/multiple-access) protocol, 
which multiplexes digital-traf- 
fic channels — that is, voice da- 
ta — into time slots. A sequence 
of six time slots makes up a 40- 
msec frame. In a full-rate traffic 
channel, a user transmits twice 
in each frame, meaning that a 
user assigned to the first time 
slot transmits again in the fourth 
time slot. By transmitting twice 
in each frame, the cell phone 
picks up EMI that looks like a 
square wave with a 20-msec, 50- 
Hz period (Figure 2). 



26 EDN I JANUARY 10, 2008 



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Figure 2 The TDMA phone standard uses radio protocols that 
result in bursts of RF at 50 Hz. You hear the demodulation 
of the signal envelope in stereos and clock radios. 



Figure 3 The GSM standard has a signal envelop with a 21 7-Hz 
frequency. Because power levels are higher and the human ear 
is more sensitive at 21 7 Hz, these phones can produce large 
interference problems. 



In contrast, the GSM (global-system- 
for- mobile ) -communication protocol 
specifies a 33-dBm transmission once 
every 4.6 msec, causing greater interfer- 
ence than the TDMA protocol, which 
transmits at 20 dBm (Figure 3) - Figures 
2 and 3 represent interference in a real- 
world system, and, in this case, the GSM 
interference is 100 mV versus 5 mV for 
the TDMA phone- The interference 
you hear in your car stereo and clock ra- 
dios is not a 900-MHz burst but a repeti- 
tive envelope of those bursts that occur 
in ICs and even in wire due to the non- 
linearity in the system. RF consultant 
James Long advises that all electronic 
devices have a transfer function that is 
a power series of the input signal That 
is. Vo„,=V,^Xkl+V,^^Xk2+V,^3xk3, 
a series that continues to infinity, with 
k representing a constant- As a result, 
many extra frequencies, including the 
demodulated baseband of the interfering 
signal, occur. Nonlinear circuits include 
those that depend on feedback to reduce 
distortion. At higher frequencies, the 
feedback effect is nonexistent, and the 
system does not suppress RFI (referenc- 
es 3, 4, and 5). 

Input-protection diodes and other 
junctions in analog ICs demodulate the 
frequencies that PCB traces and ground 
and power planes pick up, and this de- 
modulated signal appears as audio-fre- 



quency noise. At 1 GHz, the IC itself is 
not an effective antenna for typical RF 
emissions. The tiny bond wires and ca- 
pacitances are more susceptible to fre- 
quencies in the tens of gigahertz, far 
above the excitation frequencies that 
cell phones cause. Different ICs of the 
same type or from different manufac- 
turers behave differently, depending on 
variations in input capacitance or lead- 
frame inductance, but they are still sus- 
ceptible to RFI. To combat the prob- 
lem. National Semiconductor designed 
the LMV851 op amp to reject RFI. The 
company has devised the FMIRR (EMI- 
rejection-ratio) figure of merit that 
quantifies how well various pins of the 
IC reject RFI (Reference 6). 

FET and CMOS op-amp input struc- 



3.5k 




3.5k 

Figure 4 The input pins of this op amp 
have series resistors and large capaci- 
tive-clamp diodes to protect it from 
ESD. An added benefit is that the part 
is more immune to RFI (courtesy Maxim 
Integrated Products). 



tures are less prone to demodulation ef- 
fects than bipolar amplifiers are. Still, 
Kumen Blake, principal applications en- 
gineer at Microchip Technology, points 
out that CMOS parts can detect RF if 
you drive the inputs hard enough. "Fven 
CMOS will reverse-bias and create a 
transistor junction [under RF radia- 
tion]," he says. "Any op amp can convert 
RF or microwave energy into a dc signal. 
Many customers don't understand what 
symptoms they will see if they have an 
EMI problem. A dc shift can be a symp- 
tom. A change in power level means 
there's a good chance that RFI caused 
some oscillation. Another symptom is 
distortion of the signal, whether the fre- 
quency changes or whether harmonic 
distortion appears. The worst symptom 
is erratic behavior: The circuit just does 
not work right all the time." 

Some ICs use the resistance of the in- 
put structure to decouple the RF from 
inside the amplifier. Even a small input 
resistance can work with the stray ca- 
pacitance of the amplifier's ESD (elec- 
trostatic-discharge ) -protection diodes 
and other structures to effectively bypass 
the RF to ground. For example, Maxim 
uses this technique to provide ESD pro- 
tection on the LMX324 op amp and to 
provide RFI immunity (Figure 4)- The 
downside is that the resistors limit band- 
width and slightly reduce phase margin. 



28 EDN I JANUARY 10, 2008 




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Figure 5 This copper-clad board has two 
antennas soldered to opposite sides of 
the same ground plane. When you oper- 
ate circuits with fast edges on the board, 
the antennas radiate significant amounts 
of RF, even though they galvanically con- 
nect (courtesy Glen Dash). 

A ground or power plane has more 
than enough impedance to cause RFI 
reception or transmission through the 
wires that attach to the plane- You can- 
not assume that a 20X20-cm PCB with 
a ring'Style ground plane is equipoten- 
tial — that is, that every point in the 
plane is at the same potential (Refer- 
ence 7)- Glen Dash, the author of nu- 
merous papers on the laws and standards 
applicable to electronic equipment, sol- 
dered two antennas onto the sides of a 
copper-clad PCB and produced a sig- 
nificant amount of EMI by misrouting 
the digital chips on the board, causing 
large, fast-changing currents (Figure 5)- 
Experienced engineers looking at the 
telescopic antennas soldered to a com- 
mon plane would think that the system 
would not radiate RF, but they would be 
wrong. 

RF-SUSCEPTIBILITY RULES 

To understand the theory behind 
RF susceptibility, you need to know 
three general design rules: Low imped- 
ance is preferable to high impedance, 
small loop areas are preferable to large 
ones, and short wires are preferable to 
long ones. Some engineers believe that, 
when everything else fails to solve the 
problem, they must put the system into 
a shielded enclosure, but this option is 
costly and often impractical. "If design- 
ers want to avoid that expense, they 
have to do a good PCB layout," says Mi- 



AT EDN.COM 



[±] Gotowww.edn.com/080110df 
and click on Feedback Loop to post 
a comment on this articie. 



crochip's Bible. Consider that a wire in 
space is an antenna. If the wire's connec- 
tion to ground is a l-MH resistor, then 
the wire's voltage will vary more widely 
than if its connection to ground is a 511 
resistor. Gaussian law dictates routing 
two signal-carrying wires close together 
rather than in a big loop because using 
bigger loops means that the wire will 
pick up more voltage for a given RF-field 
strength. An antenna also works better 
when it is the same length as the wave- 
length of the RF field. A 1 -cm wire with 
one side that attaches to earth ground 
has a OV signal all along its length for 
frequencies of less than 1 GHz. At 900 
MFlz, a 3-in.-long wire becomes a quar- 
ter-wave antenna. Even an eight- wave 
antenna can bring significant RF energy 
into your systems. These facts highlight 
the importance of using short traces and 
tight layouts. The following rules detail 
ways that you can minimize both sus- 
ceptibility and RF emissions: 

• Attach all cables to ground, the 
power plane, or both at the same 
point. 

• Connect the sensor ground near 
where the sensor wire connects to the 
input chip. 

• Run sensor wires next to each other 
as pairs, even if one side of the sensor is 
ground or power. This approach ensures 
that common-mode interference does 
not become single-ended noise that an 
amplifier cannot reject. 

• Route the sensor wires between the 
ground and power planes, and arrange 
the decoupling capacitors in a uniform 
pattern across the planes. 

• Keep the circuit's impedances as 
low as possible within the limits of the 
components' power dissipation and the 
product's power consumption. 

• Lay out the circuit using as little 
space as possible and the smallest com- 
ponents possible within the limits of 
manufacturability and power dissipa- 
tion. 

• Keep a uniform ground plane, and 
use discipline in placement and routing 
to ensure that digital noise stays outside 
the analog circuits. 

• Reference power-supply circuits 
with large ac circulating currents to a 
topside copper pour, and then tie them 
to the ground plane at the output ca- 
pacitor's common terminal. 



30 EDN I JANUARY 10, 2008 








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are trademarks of National Instruments. Other product and 
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respective companies. 2006-8345-301-1 01 -D 



FOR MORE 


1 N FOR M ATI ON 


Aerielle 


Intel 


www.aerielle.com 


www.intel.com 


Amp Inc (now, Tyco 


James Long 


Electronics) 


www.analog-rf.com 


www.tycoelectronics. 


Kemet Electronics 


com/conn ponents 


Corp 


Analog Devices 


www. kemet.com 


www.analog.com 


Maxim Integrated 


Banner Engineering 


Products 


www. banner 


www.maxim-ic.com 


engineering. com 




Cisco Systems 


www.microchip.com 


www.cisco.com 


National 


Ford Motor Co 


Semiconductor 


www.ford.com 


www.national.com 


Glen Dash 


Palm 


http://glendash.conn 


www.palm.com 


Honda 




www.honda.com 





• Create a filtered power supply for 
each IC that requires it. Any power 
plane measuring larger than 1 in. is sus- 
ceptible to RFI. 

• Send low- impedance signals over 
any long cable runs. 

• Run signals in a stripline between 
two planes. 

• Use differential signals that don't 
depend on ground or power if possible. 

• Use 100-pF capacitors to filter out 
RE The self- inductance of a 0.1-|jlF ca- 
pacitor makes it useless at RFs. Use the 
manufacturer-supplied impedance chart 
to ensure that the capacitor you select 
has low impedance at the frequencies 
you want to suppress (Reference 8). 
The layout can have footprints for low- 
value capacitors between op- amp input 
pins, on signal-path power pins, and on 
other sensitive nodes. 

The art of analog design is knowing 
how to make trade-offs to achieve the 
desired result. Many designers do a ba- 
sic debugging of their PCBs with the 
signal layers on the outside. After meet- 
ing the fundamental requirements, they 
then make the prototype board with the 
power and ground planes on the out- 
side. This approach puts all the long 
traces that might radiate or be suscep- 
tible to EMI into a gaussian cage that 
the outer layers form. You can stitch vias 
along the edges and to separate areas. 
The vias can connect two outer ground 
planes on a six-layer board and can feed 
to decoupling capacitors on a four-layer 
board on which power is one of the out- 
side planes. A tight, low-impedance lay- 
out with careful thought about how the 
signals tie into the digital system takes 



considerable work, but this work is es- 
sential to ensure that a system has good 
RFI and EMI immunity. 

If you can do nothing to eliminate the 
source of the RFI, you must ensure that 
as little of it as possible couples into your 
circuits. After that step, judicious choice 
and diligent characterization of the ICs 
you pick for the design can improve RFI 
immunity. EDN 

REFERENCES 

9 Rako, Paul, "Measuring nanoam- 
peres," EDN, April 26, 2007, pg 42, 
www.edn.com/article/CA6434367. 
3 Rako, Paul, "Circulating currents: 
The warnings are out," EDA/, Sept 28, 
2007, pg 50, www.edn. com/article/ 
CA6372822. 

a White, Don, The EMC Desk Refer- 
ence Encyclopedia, ISBN 0-932263- 
49-6, Don White Consultants Inc, June 
1996. 

9 Hare, Ed, and Michelle Bloom, Edi- 
tors, The ARRL RFI Handbook; Practi- 
cal Cures for Radio Frequency Inter- 
ference: First Edition, ISBN-13: 978- 
0872596832, American Radio Relay 
League, February 1999. 
a Gerke, Daryl, and Bill Kimmel, EDN 
Designers Guide to Electromag- 
netic Compatibility ISBN-13: 978- 
0750676540, Newnes, August 1999. 
9 De Wagt, Gerrit, and Arie van Sta- 
veren, "A Specification for EMI Hard- 
ened Operational Amplifiers, Applica- 
tion Note 1 698," National Semiconduc- 
tor, September 2007, www.national. 
com/an/AN/AN-1 698.pdf. 
m Dash, Glen, "EMI: Why Digital De- 
vices Radiate," http://glendash.com/ 
Dash_of_EMC/Why_Devices_Radiate/ 
Why_Devices_Radiate.pdf. 
[3 "Application Notes for Multilayer Cer- 
amic Capacitors," pg 46, Kemet Elec- 
tronics Corp, www.kemet.com/kemet/ 
web/homepage/kechome.nsf/weben/ 
6FB56FCB5EBB9053CA257 
0A5001 6091 3/$file/F31 01 ECerLd 
PerfChar.pdf. 



You can reach 
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ANALOG edge- 

Noise Figure Analysis - Fully Differential Amplifier 



Application Note AN-1719 



Fully Differential Feedback Amplifiers (FDA) such as 
National's LMH6550, LMH6551, and LMH6552 are 
used to provide balanced low-distortion amplification 
and level shifting to wide bandwidth differential signals. 
A simplified conceptual diagram of an FDA is shown in 
Figure 1 where two forward paths amplify the two com- 
plementary halves of the differential signal. A separate 
common mode feedback circuit controlled by the V^^^ 
control input sets the output common mode voltage 
independent of the input common mode, as well as 
forcing the and Op outputs to be equal in magnitude 
and opposite in phase. 




Rf = Rf1 =Rf2 
Rg = Rg1 = Rg2 



Figure 1: Simplified Conceptual Diagram of Fully Differential Amplifier 

The LMH6552 is a 1.5 GHz device which allows opera- 
tion at gains greater than unity with exceptional gain 
flatness without sacrificing bandwidth. With 450 MHz 
of 0. 1 dB unity gain flatness the device is ideally suited to 
driving a range of 8- to 1 4-bit high-speed ADCs. 

In designing an FDA to drive an ADC it is required to 
ensure that the FDA does not degrade the ADC's Signal- 
to-Noise and Distortion (SINAD) performance. A key 
element of this analysis is determining and optimizing the 
noise performance of the FDA. Voltage Feedback (VFB) 
FDAs have historically been constrained to operating at 
low gain due to their poor noise performance at higher 
gains. The LMH6552 CFB architecture overcomes this 
constraint, delivering a noise advantage as well as a gain 
bandwidth advantage over alternative VFB devices. 

Output Noise Calculation for Fully Differential Amplifiers 

A general purpose FDA noise model is shown in Figure 2. 
I^p and I^^ are the input-referred noise currents for the 
FDA's positive and negative input terminals respectively. 



Mike Ewer, Application Engineer 
Robert Malone, Design Engineer 



Rs Rgi 
|-AAAr-Q 




Figure 2: Fully Differential Amplifier Noise Model 

and V^ is its input referred noise voltage. Included in the 
model are noise sources associated with resistive elements 
in the feedback and source termination networks. 



The total output referred noise density V^^ in nV/vHz, 
is calculated by taking the root square sum (rss) of the 
output referred noise from each source in the model. 
Separate the equation for V^^ into two components; the 
first being due to the input referred noise from the FDA, 
^NOFDA' second due to thermal noise from the 

resistive feedback network, V,,^^„. 



+ 1/. 



CMRR and balance error are key. The differential feed- 
back network is balanced by selecting Rp^^R^^^Rp and 
R^^=R^2^Rg' both positive and negative feedback 
factors are matched and symmetric. Replacing RS and 
RT with their Thevenin equivalent source resistance, 
Rg^p^=R^| |Rj^ the FDA and feedback network output noise 
densities are: 



Equation 1 



6f 



Na t ion a I 

Semiconductor 

The Sight & Sound of Information 



Equation 2 



National Semiconductor 

2900 Semiconductor Drive 
Santa Clara, CA 95051 
1 800 272 9959 

Mailing Address: 

POBox 58090 

Santa Clara, CA 95052 



Na t tonal 

Semiconductor 

The Sight & Sound of Information 



V^nJ = 4kTl2R,)+ 4kTl2Rjl + 4kTRs( f^sj-^ 



--Rr + 



The impact of external resis- 
tor noise is determined by the 
differential feedback topology (Equation 2), 
regardless of whether a CFB or VFB FDA is 
chosen. However, the influence of the FDA 
on total output noise density (Equation 1) 
can largely be influenced by the choice of 
FDA architecture. 

In a VFB FDA, the differential input imped- 
ance is very high (usually hundreds of ¥£1 
to several MH), and noise sources internal 
to the FDA will tend to refer to the input as 
voltages. Consequently, input referred noise 
currents will be quite small, on the order of 
a few pA/VHz, and will only contribute a 
significant portion of the total output noise 
when RF is large, which is usually not the case. 
Note that the gain term for in Equation 1 is 
simply the reciprocal of the equivalent 
feedback factor P^^^ which is related to the 
differential amplifier s closed loop gain G by: 



G = 



In other words, operating a VFB FDA at 
high values of gain is necessarily accompa- 
nied by a proportional increase in output 
noise density, and can lead to degradation of 
overall noise performance when the FDA is 
a significant source of noise in a system. 

A very different result arises when a CFB 
FDA, such as the LMH6552, is considered. 
Here the differential input stage is essentially 
a current controlled current source, with 
ideally zero differential-input impedance. 
As a consequence, noise sources internal to 
the amplifier tend to refer to the inputs as 
currents, rather than as voltages, and the 
total FDA output noise will be dominated 
by the sum of input noise currents I^p and 
I^^ multiplied by the feedback resistor 
squared. 

Unlike a VFB FDA, the output noise of 
a CFB FDA depends on the value of Rp 
rather than on the amplifier's gain. Hence, 
increasing the gain by reducing does 
not appreciably degrade the noise perfor- 
mance of the circuit. This is an extremely 
important result and highlights one of the 



key advantages of using a CFB FDA over a 
VFB FDA in differential signaling applications. 

Noise Figure Calculation for Fully 
Differential Amplifiers 

Noise performance is often described in 
terms of the system noise figure, which is 
10 log of the signal-to-noise ratio at the 
system's input, divided by the signal to noise 
ratio at its output. Noise figure is a quanti- 
tative measure of how much noise is added 
to a given signal as it propagates through a 
processing chain. For amplifiers it can be 
conveniently expressed as the ratio of output 
noise density to source noise density by the 
following equation. 

The product GD^ is the voltage gain of 
the amplifier from V^ to V^ including the 
signal attenuation of the input termination 
network. 

To highlight the difference in noise perfor- 
mance between CFB and VFB FDAs, the 
application circuit of Figure 3 is used to 
calculate the output noise spectral density 
and noise figure at various values of gain 
using both the LMH6552 (CFB) and 
LMH6550 (VFB) FDAs in a lOOn system. 
R^ is held at 30 IH and the termination 
resistor R^, is adjusted at each value of gain to 
maintain a lOOQ differentially terminated 
input. The results are presented graphically 
in Figure 3. 





- — . 






LMH6 


550 (VFE 


) 



















3 










































\ / 
















LMH6 


552 (CFE 


) 









15 I 
> 

10 "i" 



Figure 3: Noise Figure and Voltage Noise Spectral 
Density vs Gain 

Conclusion 

The choice between a current or voltage mode 
FDA may depend on many factors and will 
ultimately come down to which amplifier 
works better within a given system specifi- 
cation. Where low-noise, wide-bandwidth 
applications require the FDA to be 
configured for larger than unity gain CFB 
FDAs can offer an elegant solution. 

For Additional Design Information 



edge.national.com 



© National Semiconductor Corporation, 2007. National Semiconductor and are registered trademarks and Analog Edge is a service mark of 
National Semiconductor Corporation. All other brand or product names are trademarks or registered trademarks of their respective holders. 



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COMMUNICATIONS 
CENTRIC TEST GEAR 

SHARPENS 

SYMBOL 

RECOGNITION 

BY MAURY WRIGHT • EDITORIAL DIRECTOR 

DESIGNERS PURSUE NEXT-GENERATION WIRELESS 
DEVELOPMENTS WITH MODULATION-AWARE TEST 
TOOLS, THOUGH EVOLVING STANDARDS PRESENT 
PROBLEMS FROM THE PHY TO THE DATA LAYERS. 

Designers working on wireless systems face a moving prob- 
lem: In applications including 3G/4G cellular, WiMax, 
Wi-Fi, and UWB (ultrawideband), there is a constant 
progression to new, more complex standards. The devel- 
opment work — especially at the chip level — happens 
concurrently with standards development. Test-equip- 
ment companies are still finding ways to deliver 
products that allow development to proceed. The 
test tools increasingly include standards-spe- 
cific capabilities at the PHY (physical) and higher layers of 
the network stack. A look at some sample tools and usage 
scenarios may help your next design project — whether 
you are working on one of these wireless standards, 
on a custom project in the ISM (industrial/scien- 
tific/medical) bands, or on a wired connection. 



36 EDN I JANUARY 10, 2008 






From the test-equipment manufacturers' perspective, the 
challenge centers on anticipating the market- Jennifer Stark, 
WiMax program manager at Agilent, points to three constant 
trends- "The technology is moving to higher frequencies, wid- 
er bands, and more complex modulation schemes," she says. 

Clearly, the more complex modulation schemes offer the 
stiffer tests, as the most effective test gear includes the abil- 
ity to handle the modulation schemes in real time- The in- 
credible amount of digital-processing power readily available 
in baseband ICs makes the advances in modulation possible- 
But what you can accomplish in a baseband IC provides no 
comfort to the test-equipment company. With standards being 
moving targets, the test vendors use a combination of DSPs, 
FPGAs, and software to essentially model the transmitting 



and receiving ends of a communications link. Indeed, most 
of the test companies agree that they must virtually overde- 
sign every instrument to allow a margin for use with emerging 
standards. 

Justin Panzer, manager of product marketing at Rohde & 
Schwarz, states, "In a design environment, engineers use test 
equipment to try to simulate a real- world environment." But 
again, how do you simulate a technology in flux? Panzer points 
out that standards typically build upon one another. He uses 
the emerging LTE (long- term- emulation) standard as an ex- 
ample. The 3GPP (Third Generation Partnership Project) is 
developing LTE as a 3.5 or a 4G follow-on technology in the 
GSM (global-system-for- mobile) -communications family of 
cellular standards. "You see a lot of similarities to WiMax [in 



JANUARY 10, 2008 | EDN 37 



LTE]," says Panzer. It appears that LTE 
will employ the OFDM (orthogonal- 
frequency-division-multiplexing) and 
OFDM A (OFDM'access)'modulation 
schemes that WiMax uses, but the fre- 
quency bands and channels differ. Most 
observers believe that LTE deployment 
is at least two years away, but chip de- 
signs are well under way (Reference 1). 

MILITARY SCHEMES 

"Most things that are tried in the com- 
mercial space have been tried in the mil- 
itary space," says Agilent's Stark. She 
claims that military LMDS (locaLmuL 
tipoint'distribution-service) and MMDS 
(multichanneLmultipoint'distribution- 
service) systems were forerunners of the 
fixed flavor of WiMax. But, Stark adds, 
"Mobile WiMax is an enormous build on 
fixed WiMax." Mobile WiMax requires 
that base stations hand off calls as the cli- 
ent moves. But a Mobile WiMax system 
can't rely on channel assumptions that 
develop over time with a fixed client. 

All of the test companies participate 
at some level in the standards bodies 
and therefore enjoy some visibility in 
shaping the direction of and even hav- 
ing influence over some decisions. "You 
can see two, three, or even four years 
out sometimes in the standards bodies," 
says Graham Celine, senior director of 
marketing at Azimuth Systems. He also 
notes that disruptive standards can arise. 
For example, on a standard for the 700- 
MHz spectrum, the FCC (Federal Com- 
munications Commission) will hold an 
auction in the United States in February. 
TV broadcasters are about to vacate this 
spectrum. The highest bidder could for 
years potentially influence the course of 
US broadband history. "Until 
someone wins that auction, no- 
body knows," says Celine. 

As an example of how test 
companies influence standards, 
consider a WiMax case. Anto- 
nio Policek, senior marketing 
manager in Tektronix's commu- 
nications-business unit, claims 
that Tektronix influenced the 
inclusion of a metering port in 
base-station designs. The com- 
pany implements the metering 
port at the output of the base- 
band stage and allows a pro- 
tocol analyzer to gather data 
without having an RF receiver 



AT A GLANCE 

□ Wireless technologies push 
center frequency, bandwidth, and 
modulation complexity. 



□ Test companies scrutinize stan- 
dards bodies but must anticipate 
the market. 



□ A signal generator, a signal 
analyzer, and modulation software 
can simulate a wireless network 
in the lab. 



□ Communication-test gear allows 
you to optimize the design of criti- 
cal functions, such as the power 
amplifier. 



in the instrument. It seems that the test 
community largely seeks to ensure that 
standards are testable. "Azimuth wants 
to make sure that the test models make 
sense and can reasonably be implement- 
ed in instruments," says Celine. 

MODELING WIRELESS SYSTEMS 

It's amazing how easy it is to model 
a wireless-communication system with 
modern test equipment in a develop- 
ment lab. National Instruments, for 
instance, offers a number of RF-signal- 
generator and -analyzer products in the 
PXI ( peripheral-component- intercon- 
nect-extensions-for- instrumentation) 
and PXIe (PCI Express-extensions-for- 
instrumentation) form factors, which 
are essentially ruggedized versions of 
PCI and PCIe (PCI Express). You can 
use what's essentially a PC with these 
hardware modules installed and com- 
pletely model a cellular system using the 
company's Modulation Toolkit software 
running on the Lab View graphical-de- 
velopment environment. Figure 1 de- 




Figure 1 The Modulation Toolkit, which runs on National 
Instruments' LabView, allows designers to quickly model 
wireless systems and offers diagnostic tools, such as this 
constellation and eye diagram. 



picts a symbol constellation and 3-D eye 
diagrams from a 16-QAM (quadrature- 
amplitude-modulation) system. 

A number of case studies illustrate the 
usage of the National Instruments tools. 
For instance. Reference 2 describes a 
joint project between the University of 
Texas and Drexel University focusing 
on modeling MIMO (multiple- input/ 
multiple-output) systems. The Univer- 
sity of Texas also used LabView in 2003 
to model early WiMax systems. 

The modularity of National Instru- 
ments' products also lends the tools to 
fieldwork. Recording RF energy in the 
field is one common task designers use 
to test products in the lab. It turns out 
that it's tough to synthetically create the 
difficult environment that's a reality in 
the field. David Hall, National Instru- 
ments product manager, points out that 
you can use one of the company's VSAs 
(vector-signal analyzers) in a system 
with a hard drive to record five hours 
of a real-life environment that you can 
subsequently replay in the lab. 

PUSHING FREQUENCY 

To support testing and modeling of 
state-of-the-art wireless systems, the test 
companies must push their hardware de- 
sign and add the software-based modu- 
lation tools. Going back to the three 
dimensions in which wireless technolo- 
gies progress, Agilent's Stark notes that 
individual standards tend to push in one 
or two of the dimensions but not in all 
three. She cites examples in WiMax and 
UWB. Stark claims that UWB, the most 
prevalent WiMedia flavor, uses a rela- 
tively simple modulation scheme but rel- 
atively wide 500-MHz channels. WiMax, 
conversely, uses relatively nar- 
row 10-MHz channels but a 
complex modulation scheme. 
In both cases, the standards 
specify operation in the 5- to 
6-GHz range. 

Just the center frequency 
of a new standard can lead to 
the need for new hardware. 
Panzer of Rohde & Schwarz 
points out that the company's 
CMU200 mobile-radio tester 
can work with virtually ev- 
ery cellular standard. But it 
operates only to 3 GHz, and, 
to support WiMax, the com- 
pany had to develop the 6- 



38 EDN I JANUARY 10, 2008 



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GHz CMW500 tester. And the hard^ 
ware push doesn't stop there. To test a 
MIMO system, you would need two of 
the 6'GHz units. 

Azimuth's Celine claims that MIMO 
was a difficult technology to support in 
test tools. Azimuth has focused on Wi- 
Fi and WiMax and bet on MIMO ex- 
pertise early on as a way to differenti- 
ate the company. Celine points out that 
the channel emulator in a MIMO test 
instrument differs from the one in a tra- 
ditional instrument. "In SISO [single 
input/single output], the emulator acts 
as an interference source," says Celine. 
"In MIMO, multipath is what makes the 
technology work." 

AMP OPTIMIZES EXPERIENCE 

Some usage scenarios help illustrate 
other instrument features and design chal- 
lenges and detail just how you might use 
such an instrument in the lab. Agilent's 
Stark points at the power amplifier as a 
crucial part of an OFDMA radio design 
for a standard such as Mobile WiMax. 
She claims that a poor power-amp design 
can result in poor battery life, range, and 
data rate — attributes of a product that ul- 
timately matter a lot to the consumer. 

According to Stark, the power am- 
plifier is an issue in the Mobile WiMax 
case, because the modulation scheme 
causes the design to drive the amp in a 
nonlinear range. Moreover, the output 
signal must change erratically and has 
a high peak-to-average ratio. The client 
side of a WiMax design also has space 
and heat constraints. 

In this age of simulation and comput- 




Figure 2 By continuously performing FFTs, 
Tektronix's real-time-spectrum-analyzer 
family can catch transients as short as 24 
|jLsec, allowing the instruments to display 
overlapping-channel noise in OFDM 
systems. 



er tools, you needn't build hardware to 
start a power-amp development. You can 
design and simulate the amp using Agi- 
lent's EEsof RF-modeling EDA tool. The 
EEsof tool can feed a signal generator, 
and you can use the 89600 VSA software 
on a PC to characterize your design. The 
VSA package runs on a PC and interfac- 
es with a variety of Agilent oscilloscopes 
and signal-analyzer instruments. 

Stark offers several examples of speci- 
fications that you can characterize and 
tune in this scenario. For example, a 
spec in WiMax and cellular standards 
that's commonly called either EVM (er- 
ror-vector-magnitude) or RCE (relative- 
constellation-error) measures constel- 
lation accuracy. The power amplifier is 
one component that adds to EVM/RCE 
errors. A simple system that uses, say, 
BPSK (binary phase-shift keying) or 4- 




With a frequency range to 3.3 GHz and optionally to 6 GHz, the Rohde & Schwarz 
CMW500 mobile-radio tester supports evolving standards, such as Mobile WiMax. 






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QAM (four-phase QAM), widely spaces 
the modulation symbols, and such a sys- 
tem can tolerate a high EVM/RCE fig- 
ure. But a 64''QAM (64''phase QAM) 
system has a tight error budget, and, ac- 
cording to Stark, the amplifier's design 
often contributes a maximum of 1% of 
that error budget- 

You can perform EVM/RCE tests only 
on a working radio. But the simulation 
component in Stark's scenario allows 
testing and optimizing before commit- 
ting the design to hardware. 

AMPS POSE PROBLEMS 

Darren McCarthy, worldwide RF- 
technology-marketing manager in Tek- 
tronix's instrument-business unit, agrees 
that power- amplifier design is crucial to 
battery life and performance in wireless 
clients. McCarthy points out the need 
in an OFDM or OFDMA system to min- 
imize power leakage from one channel 
into another. The test for such an occur- 
rence is the ACPR (adjacent-channel- 
power ratio), a measurement of the lin- 
earity in a system. 

As noted, however, systems such as 
Mobile WiMax and LTE drive power 
amplifiers into nonlinear regions — nega- 
tively impacting the ACPR. Increasing- 
ly, designers are turning to techniques 
such as DPD (digital predistortion) to 
minimize ACPR. But DPD introduces a 
new problem: memory effects, which re- 
sult from electrical traits, such as source 
and load impedance and electrothermal 
coupling. 

Traditionally, designers use spectrum 
analyzers and software tools to measure 
ACPR. But McCarthy claims that such 
ACPR measurements measure average 
power and that traditional spectrum an- 
alyzers using a swept-spectrum approach 
miss transient signals, such as those that 
memory effects cause. Tektronix offers 
a family of real-time spectrum analyzers 
that continuously perform frequency-do- 
main transformation to catch memory- 
effect transients. Figure 2 shows a Tek- 
tronix digital-phosphor-spectrum display. 



MORE 



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\±\ Gotowww.edn.com/080n0cs 
and click on Feedback Loop to post 
a comment on this articie. 




The 89601 A LTE signal-analysis package 
from Agilent performs an EVM (error-vec- 
tor-magnitude) measurement on an LTE 
downlink signal, and the package displays 
a constellation diagram. 

The yellow curve indicates the maximum 
occurrence of noise in the adjacent chan- 
nels. McCarthy claims that the instru- 
ment will capture any transient that lasts 
24 fxsec or longer. 

Equally important, according to Mc- 
Carthy, is the ability to tie the occur- 
rence of a problem transient with the 
root cause. The instruments support fre- 
quency-masked triggering, and you can 
connect that trigger to a variety of oth- 
er instruments. McCarthy claims that 
you can use such a trigger to locate the 
software instruction that caused a gain 
change in the amplifier and, therefore, 
the transient. 

UWB AND OTHER STANDARDS 

Wide-area wireless technologies aren't 
the only technologies that are changing. 
And designers certainly need good test 
gear for standards such as UWB. Tek- 
tronix's McCarthy claims that standards 
such as UWB and IEEE 802.1 In offer 
special challenges because they offer 
"cognition of the environment." Such 
cognitive radios are especially impor- 
tant in standards that use the unlicensed 
spectrum, because the radios must avoid 
interfering with other transmitters. 

The UWB community has developed 
the DA A (detect-and-avoid) technique 
to ensure that the broad transmissions 
don't interfere. Europe, Japan, and some 
other regions will mandate DAA. Mc- 
Carthy points out that, although UWB's 
transmitting power is far lower than that 
of, say, a cellular transmitter, a UWB 
transmitter could still interfere with 
a cellular receiver. So, DAA basically 
specifies listening for a transmitter and 




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avoiding frequencies in which a trans- 
mitter is operating. 

Several operational modes exist for 
DAA implementations- UWB features 
three 500-MHz channels among which 
UWB radios hop- In the simplest form, 
DAA does not use one of the three 
channels when another transmitter is 
present- In many cases, however, a sys- 
tem can take a more granular approach 
to avoiding interference. Using a tone- 
nulling technique, a transmitter sim- 
ply avoids a narrow range of subcarri- 
ers within one of the 500-MHz bands. 
According to McCarthy, the Tektronix 
AWG7000 signal generator with option- 
al UWB software allows design teams to 
test all DAA modes. 

THE RUSH TO LTE 

Looking forward, all of the test ven- 
dors are working feverishly on LTE 
products in the lab and shipping some 
products, as well. Panzer from Rohde 
& Schwarz claims that the company 
early on engaged with a leading chip 
vendor and has delivered gear that can 



handle PHY and RF tests. Indeed, the 
CMW500, which also supports WiMax, 
offers LTE support. 

Azimuth's Celine maintains that the 
challenge everyone will face is effec- 
tive MIMO support. He believes that 
the MIMO experience the company has 
gained with WiMax will serve it well in 
an upcoming LTE product. "LTE is in a 
very different frequency band," he says. 
"We are redesigning the RF front end, 
but that's easier than handling a new 
modulation scheme." 

Regardless of what type of system you 
are working on, you might consider one 
common thought that all of the test ven- 
dors have voiced: Standards for technol- 
ogies such as wireless communications 
don't provide design advice. "A standard 
document defines system performance," 
says Agilent's Stark. "It doesn't tell the 
engineer how to design." You probably 
knew that fact. But it means that you 
had better thoroughly test your design 
and that wireless channels probably offer 
more unknowns than any other environ- 
ment in which you will ever work.EDN 



FOR MORE INFORMATION 



Agilent Technologies 

www.agilent.com 

Azimuth Systems 

www.azimuthsystems. 
com 

National Instruments 

www. ni. com 



Rohde & Schwarz 

www.rohde-schwarz. 
com 

Tektronix 

www.tek.com 

3GPP 

www.3gpp.org 



REFERENCES 

D Wright, Maury, "WiMax gains in mo- 
bile-broadband game, but 4G lurks," 
EDA/, March 29, 2007, pg 56, www. 
edn.com/article/CA6426878. 
B Heath, Robert W, Jr; Kapil R Dandek- 
ar; and Scott M Nettles, "UT and Drexel 
use NI PXI and LabView for wireless 
research," http://sine.ni.com/csol/cds/ 
item/vw/p/id/587/nid/1 241 00. 



You can reach 

Editorial Director 

Maury Wright 

at 1-858-748-6785 and 

mgwright@edn.com. 





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BY JIM WILLIAMS AND MARK THOREN • LINEAR TECHNOLOGY 



Novel measurement 
circuit eases battery- 
stack-cell design 

A TRANSFORMER AND DIODE ON EACH CELL ALLOWS 
ISOLATED MEASUREMENT 



Automobiles, aircraft, marine vehicles, uninter- 
ruptible power supplies, and telecom hardware 
use series'connected battery stacks. These 
stacks of individual cells may contain many 
units, potentially reaching hundreds of volts- 
In such systems, it is desirable to accurately 
determine each individual cell's voltage. Obtaining this in- 
formation in the presence of the high common-mode volt- 
age that the battery stack generates is more difficult than you 
might suppose. 

The "battery-stack problem" has been around for a long time. 
Its deceptively simple appearance masks a stubborn problem. 
Designers have tried various approaches to isolated-cell-volt- 
age measurement with varying degrees of success (see sidebar 
"Some battery-cell-measurement techniques just don't work" 
at the Web version of this article at www.edn.com/ms4255). 

Figure I's voltmeter measures a single-cell battery. Beyond 
the obvious benefits, the arrangement works because no volt- 
ages other than the single cell lie in the measurement path. 
The ground-referred voltmeter encounters only the voltage it 
is measuring. 

Figure 2's stack of series-connected cells is more complex. 
The voltmeter must switch between the cells to determine 
each cell's voltage. Additionally, the voltmeter, normally 
composed of relatively low-voltage breakdown components, 
must withstand input voltages relative to its ground terminal. 
This common-mode voltage may reach hundreds of volts in 
large series-connected battery stacks, such as those in an au- 
tomobile. Such high-voltage operation is beyond the voltage- 
breakdown capabilities of most practical semiconductor com- 
ponents, particularly if the application requires accurate meas- 
urement. The switches present similar problems. Attempts at 
implementing semiconductor-based switches encounter dif- 
ficulty due to voltage-breakdown and leakage limitations. A 
practical method is necessary that would accurately extract 
individual cells' voltages and reject common-mode voltages. 
This method should not draw any battery current and should 
be simple and economical. 

Figure 3's concept addresses these issues. To determine bat- 
tery voltage, Vg^^^gj^Y' ^ pulse excites a transformer, T^ and 
records its primary clamp voltage after settling occurs. The 
diode and the battery-voltage shunt primarily set this clamp 
voltage and similarly clamp T^'s secondary. The diode and a 



VOLTMETER 



GND- 



SINGLE-CELL 
BATTERY 



O--, 

+ 1 

O-J 

+! 



' N CELLS 



Figure 1 Common-mode voltage does 
not affect a voltmeter measuring a 
ground-referred single cell. 



VOLTMETER 



GND- 



SWITCH 
CONTROL 

GND 



BATTERY 
STACK 



Figure 2 A voltmeter measuring a cell 
in a stack undergoes increasing com- 
mon-mode voltage as the measurement 
proceeds up the stack. The switches 
and switch control also encounter high 
voltages. 



small transformer term are predictable errors, and the circuit's 
final stage substracts them out, leaving the battery voltage as 
the output. 

DETAILED CIRCUIT OPERATION 

Figure 4 details the transformer-based sampling voltmeter. 
It closely follows Figure 3 with some minor differences, which 



JANUARY 10, 2008 | EDN 47 



PULSE 
GENERATOR 



DELAYED-PULSE 
GENERATOR 



DIODE T N CELLS 




PULSE p 
GENERATOR J 



ANALOG INPUT 

SAMPLING 
VOLTMETER 

SAMPLE COMMAND 



PRIMARY 

DELAYED 
PULSE _ 



_r 



J N CELLS 



. DC POTENTIAL= 
T/S CLAMP VALUE 



DIODE-TRANSFORMER 
ERROR TERMS 
SUBTRACTION 



^BATTERY+DIODE 

AND T. ERROR 



■OOUTPUT=Vbattery 



Figure 3 A transformer-based sampling voltmeter operates independently of high common-mode voltages. The pulse generator 
periodically activates T^. The delayed pulse triggers a sampling voltmeter, capturing T^'s clamped value. Residual error terms are 
corrected in the following stage. 



this article later describes. The pulse generator produces a 10- 
|jLseC'wide event (Trace A, Figure 5) at a l-kHz repetition 
rate. The pulse generator's low- impedance output drives T^ 
through a lO-kll resistor and triggers the delayed-pulse gen- 
erator. T^'s primary, Trace B, responds by rising to a value rep- 
resenting the sum of the diode voltage and the battery voltage, 
along with a small fixed error that the transformer contributes. 
T^'s primary becomes clamped at this value. After a time, the 
delayed pulse. Trace C, generates a pulse. Trace D, closing Sp 
allowing to charge toward T^'s clamped value. After a num- 
ber of pulses, assumes a dc level identical to the voltage 
on T^'s clamped primary. A^ buffers this potential and feeds 
differential amplifier A^. A^, operating at a gain near unity, 
subtracts the diode- and transformer- error terms, resulting in a 
direct reading of battery- voltage output. 

Accuracy critically depends on transformer-clamping fidel- 
ity over temperature and clamp-voltage range. The carefully 
designed, specified transformer yields Figure 6's waveforms. 
Primary, Trace A, and secondary. Trace B, clamping details ap- 
pear at a highly expanded vertical scale. Clamping flatness is 
within millivolts; trace-center aberrations derive from S^-gate 
feedthrough. Tight transformer-clamp coupling promotes good 
performance. Circuit accuracy at 25°C is 0.05% over a to 2V 
battery range with 120 ppm/°C drift, degrading to 0.25% at a 
battery voltage of 3V. Designers of this circuit used a floating 
variable-potential battery (see sidebar "Floating-output, vari- 
able-potential battery simulator" at the Web version of this 
article at www.edn.com/ms4255). 

Several details aid circuit operation. The circuit substitutes 
the transistor's base-emitter voltage for diodes, providing more 
consistent initial matching and temperature tracking. The 
10-fxF capacitor at maintains low impedance at frequency, 
minimizing cell-voltage movement during the sampling inter- 



val. Finally, synchronously switched prevents T^'s negative- 
recovery excursion from deleteriously influencing S^'s opera- 
tion. 

This approach's advantage is that its circuitry does not en- 
counter high common-mode voltages; T^ galvanically isolates 
the circuit from common-mode potentials associated with the 
battery voltage. Thus, you can employ conventional low-volt- 
age techniques and semiconductors. 

MULTICELL VERSION 

The transformer-based method is inherently adaptable to 
the multicell-battery-stack-measurement problem. Figure 7's 
conceptual schematic shows a multicell-monitoring version. 
Each channel monitors one cell. You can read any channel 
by biasing its appropriate enable line to turn on a FET switch, 
enabling that channel's transformer. The hardware for each 
channel typically includes only a transformer, a diode-con- 
nected transistor, and a FET switch. 

AUTOMATIC CONTROL AND CALIBRATION 

This scheme suits digital techniques for automatic cali- 
bration. Figure 8 shows pulse generators, calibration chan- 
nels, and measurement channels, which feed Figure 9's PIC- 
16F876A microcontroller. As before, even though the cell 
stack may reach hundreds of volts, the transformer's galvanic 
isolation allows the signal-path components to operate at low 
voltage. The design includes an automatic calibration-circuit 
microcontroller and reset sections (Figure 9) and an auto- 
matic calibration-circuit USB interface for development only 
(Figure 10). 

A further benefit of processor-driven operation is the elimi- 
nation of Figure 4's base-emitter-voltage diode-matching re- 
quirement. In practice, engineers tested a processor-based 



48 EDN I JANUARY 10, 2008 



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board at room temperature with known voltages at all input 
terminals- They then read the channels, which furnished the 
information necessary for the processor to determine each 
channel's initial base-emitter voltage and gain. The engineers 
then stored these parameters in nonvolatile memory, permit- 
ting a one-time calibration that eliminates both base-emitter- 
voltage-mismatch- and gain-mismatch- induced errors- 
Channels 6 and 7 provide and 1.25V reference voltages, 
representing cell-voltage extremes- The room- temperature 
values reside in nonvolatile memory. As temperature changes 
occur, you use readings from channels 6 and 7 to calculate a 
change in offset and a change in gain that you apply to the six 
measurement channels- The calibration continues as temper- 



A=5V/DIV 



B=2V/DIV 



C=5V/DIV 



D=5V/DIV 




5V 



5V, 1 ijlSEC, 1 kHz 



74HC04 



OUT IN 
LT1761-5 
GND 



h 74HC04 



\ 74HC04 



ye 74HC04 



\ 74HC04 



T 



12V,K 



2 jjlSEC/DIV 

Figure 5 Figure 4's waveforms include the pulse-generator 
input (Trace A), the primary (Trace B), the 74HC1 23's Q2 
delay-time output (Trace C), and S/s control input (Trace D). 
Timing ensures that sampling occurs when T^ settles in the 
clamped state. 



PULSE 
GENERATOR 



MOk ^-^DIODE 



SAMPLING VOLTMETER 
(SAMPLE AND HOLD) 





12V 



5VO- 



► 7.5k 

; 1% 

lOOpF 

tiL 



5V 



0.001 |jlF 12V 





Vcc RC1 01 QI Q2 0L2 B2 A2 
74H0123 
A1 B1 0L1 C2 RC2 G 




22k 



Q ERROR SUBTRAOTION 

2N3904 



511 



DELAYED- 
PULSE 
GENERATOR 




NOTE: T. IS A PULSE ENGINEERING PA21 OONL OR PA21 01 NL. 



Figure 4 This transformer-fed sampling voltmeter closely follows Figure 3's concept. Error-subtraction terms include Qg's compensa- 
tion for and resistor/gain corrections for errors in T^'s clamping action. Transistors Q^, Q^, and Q3 replace diodes for more consis- 
tent matching. Q2 prevents T^'s negative-recovery excursion from influencing S^. 



50 EDN I JANUARY 10, 2008 




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A=2 mV/DIV 
ON 2.2V STEP 




B=2 mV/DIV 
ON 2.2V STEP 



2 uSEC/DIV 

Figure 6 T^'s primary (Trace A) and secondary (Trace B) clamp- 
ing details have a highly expanded vertical scale showing the 
primary's and secondary's clamping flatness to be within mil- 
livolts. The trace aberrations at the center derive from S^'s gate 
feedthrough. 



TO 

SAMPLE- o 
AND-HOLD 
AMPLIFIER 




BATTERY 
T STACK 



Figure 7 Adding 
enable lines and 
transistor switches 
facilitates multiple 
channels. 



N SECTIONS 



OFFSET 

2N3904 



PROCESSOR- 
DATA BUS 




RBI /LATCH O— >CLK 



-±- 5VO— |IN OUT|-' 

LT1 790-1.25 

G G 



(a) 



2.2 plF 

ADC_CHO C 
ADC_CH1 C 
ADC_CH2 C 
ADC_CH3 C 
ADC_CH4 C 
ADC_CH5 C 
ADC_CH6 C 
ADC_CH7C 



10 |xF 



vcc 
LTC1 867 

Vref 

CHO 
CHI 
CH2 
CHS 
CH4 
CH5 
CH6 
CH7 

REFCOMP 
GND 



lO.I fiF_^ 1 ixF SIX IDENTICAL CHANNELS 



> RB5/CS 

> RC3/SCK 

> RC5/M0SI 

> RC4/MIS0 



74HC14 , 



PROCESSOR I 
SPI BUS 



ENX 

R BO/ EXCITATION 



74HC00 




(b) 



Figure 8 ADC-calibration channels eliminate base-emitter-voltage-matching requirements (a) and compensate for temperature- 
dependent errors (b). 



52 EDN I JANUARY 10, 2008 



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«4 

1-1 k 




ature varies because each channel's — 2-mV/°C base-emitter- 
voltage-drift slopes are nearly identical. Similarly, gain errors 
from channel to channel are nearly identical. 

Because you are continuously calibrating the gain and offset, 
the gain and offset of the LTC1867 drop out of the equation. 
The only points that must be accurate are the 1.25V refer- 
ence voltage, which an LTl 790-1. 25 IC provides, and the OV 
measurement, which is easy: Just short the Channel 6 inputs 
together. The LTC1867 internally amplifies its internal 2.5V 
reference to 4-096V at the reference-comparator pin, which 
sets the full scale of the ADC: 4-096V for unipolar mode and 
± 2.048V in bipolar mode. Thus, the absolute maximum cell 
voltage that you can measure is 3.396V. And, because the off- 
set measurement is nominally 0.7V at the ADC input, it is 
never in danger of clamping at OV. A OV reading results if the 
LTC1867 has a negative offset and the input voltage is any 
positive voltage less than or equal to the offset. Accuracy of 
the processor-driven circuit is 1 mV over a to 2V input range 
at 25 °C. Drift drops to less than 50 ppm/°C — almost three 
times lower than that in in Figure 4. 

The complete firmware code. Listing 1, is available with 



the Web version of this article at www.edn.com/ms4255. 
The code for this circuit is a good starting point for an actual 
product. Data appears on a PC screen through an FTDI (Fu- 
ture Technology Devices International, www.ftdichip.com) 
FT242B USB-interface IC. The PC has FTDI's virtual-com- 
munications-port drivers installed, allowing control through 
any terminal program. Data for all channels continuously ap- 
pears on the terminal, and simple text commands control pro- 
gram operation. 

A timer interrupt occurs 1000 times per second. It controls 
the pulse generators and ADC and stores the ADC readings 
in an array that you can read at any time. Thus, if the main 
program is reading the buffer, the most out of date any read- 
ing is is 1 msec. 

The software also includes automatic calibration routines. 
Two functions store a zero reading and a full-scale reading for 
all channels, including the calibration voltages you apply to 
channels 6 and 7, to nonvolatile memory. You subsequently 
use these functions to calibrate out the initial gain and offset 
errors, as well as the temperature-dependent errors. The entire 
procedure is to apply OV to all inputs and issue a command to 



0.1 |jlF 



PIC1 6LF876A-I/PT 



:0.1 |jlF 



0.1 |jlF . 



V+ OUT 
GND 
SET DIV 



22.1 



20 MHz 



0.1 |jlF 
0.1 |jlF 



SUPERVISOR 



5V 
O 



EN 
DIS 



14k 
102k 



2200 pF VcC25 VcCsVcCA 

<i — I I RT RST 

100 pF LTC1726EMS8-2.5 



WT WDI 
GND 




PRE Vcc 
Q D 



Q CLK<- 
CLR GND 



PRE 
Q 



Q CLK<- 
CLR GND 



5 USB RD 
CLK# 



RESET VIA USB 



22.1 
-^NAAr- 



SYSTEM RST# 



Vdd Vdd 
RA5 
RA4 

>OSCIN RA3 
RA2 
RA1 
RAO 
PGD/RB7 
PGC/RB6 
RB5 
RB4 
RB3 
>OSCOUT RB2 
RBI 
I NT/ R BO 
RX/RC7 
TX/RC6 
MOSI/RC5 
MISO/RC4 
SCK/RC3 
RC2 
LR RC1 
RCO 

RD7 
RD6 
RD5 
RD4 
RD3 
RD2 
RD1 
RDO 

Vnn 



RE2 
RE1 
REO 



RA4/ISO PWRSHDN^ 



USB WRCLK . 



USB RDCLK#^ 



IN-CIRCUIT- 
PROGRAMMING 
PORT 



O O 
O O 
O O 



Q 0|— j 



RBO/EXCITATION , 



PROCESSOR- 
DATA BUS 



■ COMMAND 
^475 



5V 5V 5V 

J J J 

^475 ^475 ^47 

RCs RCi RCo 




O10M 



Figure 9 The design includes an automatic-calibration-circuit microcontroller and reset sections. 



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PROCESSOR- 
DATA BUS 



DB7 O 

DB6 O- 

DBS O- 

DB4 O- 

DB3 O- 

DB2 O- 
DB1 
DBO 



0.1 |jlF 

— 0.1 |jlF 

H 

0.1 |jlF 

H 



5V 

Q 



USB 

5V 



o- 



USB_RD_CLK# O- 

USB_WR_CLK O 

USB_TXE# O- 

USB_RXF# O 



MOk 

22.1 

— 48 MHz O ^AA^■ 

USB, 

5V 



10k 
10k> 

5B 1 



O-C PWREN# 



USB 
5V 

0.1 |jlF 9 LTC6905CS5-96 




0.1 |jlF— ^ 



6.3V 



AVccVcc VccV< 
D7 
D6 
D5 
D4 
D3 
D2 
D1 
DO 



CC VCC vcc VCCIO 

USBDM 

USBDP 
FT245BM 

RSTOUT# 



0-w 



-0 RD# 

WR 
C TXE# 
C RXF# 

l-l XTIN 

XTOUT 

SW/WU 



EECS 
EESK 

EEDATA 
3.3VniiT 



TEST 



RST# 

AGND GND GND 



22.1k 
-JSAAr- 

22.1k 



USB 

5V 

o 

0.1 |jlF 



'A7 pF : 



!47 pF 



- USB-B 
CONNECTOR 



jj^^P^l^iok 



EE 



EN I 

Disr 



0.1 |jlF 



USB 

5V 

I ? 10k 
I fi 1 

I Vcc DO ^— 1 

CS Vss I 



Vcc DO 
CS Vss 
CLK Dl 



2.21k 



93LC46BT-1/OT 



Figure 1 The design includes an automatic-calibration-circuit USB interface for development only. 













































^CLFIMP 






















1,,,, 








. . . 1 ' 














IpULSE 


.... 




















|mosi 
























n r 














Ics 
IsCK 


















O. 


J 


r 






IMISO 

















OUTPUT_D 
(LATCH WORD [J]); 



OUTPUT_HIGH (LATCH); 
OUTPUT_LOW (LATCH) 
OUTPUT_LOW (CS) 



HIGHBYTE= 
SPLREAD 
(LTC1867- 
CONFIG [J]); 

OUTPUT_HIGH 
(EXCITATION) 



DELAY_US 
(DELAY); 

LOWBYTE= 
SPLREAD (0); 



OUTPUT_ 
HIGH 

(CS); 



OUTPUT_LOW 
(EXCITATION); 



Figure 1 1 The interrupt-service routine monitors digital signals, 
excitation pulse, and clamp voltage at the ADC input along with 
the C code that performs these operations. 























p 






















ICLFIMP 










































Ipulse 






















Ilrtch 






I 


f 


















1 




1 
































1 1 


1 1 


1 1 


1 1 


TT 


1 1 


TT 


1 1 














-L_L 



i = 0; i=1; i = 2; i = 3; i = 4; i = 5; i = 6; i = 7; 
Figure 1 2 The interrupt-service routine scans eight channels. 



store the zero calibration, then apply L25V to all inputs and 
issue a command to store the full-scale calibration. Note that 
this procedure is no more complicated than a basic perform- 
ance test that would be part of any manufacturing process. 
The 1.25V factory-calibration source can be from a voltage 
calibrator or from a selected LT 1790- 1.25 that you keep at a 
stable temperature. 
The software also includes a digital filter for testing purpos- 



56 EDN I JANUARY 10, 2008 





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es- The filter is a simple exponential IIR (infinite-impulse-re- 
sponse) filter with a constant of OA. This filter reduces the 
noise in the readings by a factor of the square root of 10. 

MEASUREMENT DETAILS 

To take a reading from a given channel, the processor must 
apply the excitation to the transformer, wait for the voltage 
signal to settle out, take a reading with the ADC, and then re- 
move the excitation. To perform these tasks, an interrupt-ser- 
vice routine occurs once every millisecond. For the details, see 
Listing 1 at www.edn.com/ms4255. Figure 11 shows the digi- 
tal signals, excitation pulse, and clamp voltage at the ADC 
input along with the C code that performs these operations. 
Loading a 3'bit byte high into the 74HC574 latch enables in- 
dividual channels. 

Note that you apply the excitation after 8 bits of the 



LTC1867 data are read out. This situation is perfectly accept- 
able, because no conversion is taking place, and all of the data 
in the LTC1867 output register is static. Depending on the 
timing of the processor you use, you can apply excitation be- 
fore reading any data, in the middle of reading data, or after 
reading the data but before initiating a conversion. If the se- 
rial clock is slow — 1 MHz, for instance — applying excitation 
before reading any data would result in the excitation being 
applied for 16 fxsec, which is too long. The only constraints 
are that the voltage at the ADC input must have enough time 
to settle properly and that you do not leave the excitation on 
for too long. Figure 1 2 shows the same signals over the entire 
interrupt-service routine. Similar analog signals are at each 
transformer and the other LTC1867 inputs. 

Many ways exist to add channels to this circuit. Figure 13 
shows a 64'channel concept that decodes the 64 channels in- 



74HC4051 




LTC1867_CH7 



LTC1 867 
INPUTS 



LTC1867 CHO 
O 



CHANNELO- 
SELECTIONO- 



WITHIN BANKO- 



74HC138N 



BANK SELECTION O- 
(FOLLOWS SELECTED O- 
LTC1867 INPUT) O- 



EXCITATION 
O— 



G1 

G2A 

G2B 



YO 0— 
Y1 0— O 
Y2 0— O J 
Y3 0— O J 
Y4 0— O J 
Y5 0— O * 
Y6 0— O 
Y7 0— 



EXCITATION 
O— 



A 

B Y1 b— O 

C Y2 0—0 

Y3 0—0 
Y4 0—0 
G1 Y5 0—0 

G2A Y6 0—0 
G2B Y7 0— 



OCH63+ 
10 |xF 
OCH63~ 



SIX ADDITIONAL 
CIRCUITS PER 
BANK 
2N3904 

OCH56+ 



; SIX ADDITIONAL 
; BANKS 



74HC4051 




X 


XO 




XI 




X2 




X3 


A 


X4 


B 


X5 


C 


X6 


INH 


X7 



EXCITATION 
O— 



Y1 0— O 
Y2 0— O 
Y3 0— O 
Y4 0— O 
Y5 0— O 
0| G2A Y6 0— O 
G2B Y7 0— 



G1 



10k 




O CH7"'" 
10 |xF 
O CH7" 




SIX ADDITIONAL 
CIRCUITS PER 
BANK 
2N3904 

OCHO+1 



Figure 1 3 A 64-channel concept decodes the 64 channels into eight banks of eight channels using 74HC1 38 address 
decoders. 



58 EDN I JANUARY 10, 2008 



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to eight banks of eight channels using 74HC138 address de- 
coders- The selected bank corresponds to one LTC1867 input 
that is programmed through the SPL Some S-to-l 74HC4051 
analog switches perform the additional analog multiplexing. 
A single 74HC4051 feeding each LTC1867 input gives 64 in- 
puts- The LTC1867, rather than a single-channel ADC, is still 
a great choice in high-channel-count applications, because it 
is good idea to break up multiplexer trees into several stages to 
minimize total channel capacitance- The LTC1867 takes care 
of the last stage. With a maximum sample rate of 200k sam- 
ples/sec, it can digitize as many as 200 channels at the maxi- 
mum Ik-sample/sec limitation of the sense transformer. That's 
a lot of batteries.EON 



REFERENCES 

n Williams, Jim, "Transformers and optocouplers implement 
isolation techniques," EDA/, Jan 6, 1 982, pg 11 5. _ 
3 Williams, Jim, "Isolated Temperature Sensor," 
LT1 98A Data Sheet, Linear Technology Corp, 
1983. 

3 Dobkin, RC, "Isolated Temperature Sensor," 
LM135 Data Sheet, National Semiconductor 
Corp, 1 978. 

a Williams, Jim, "Isolation Techniques for Signal 
Conditioning," National Semiconductor Corp, 



Application Note 298, May 1982. 

aSheingold, DH, Transducer Interfacing Handbook, Analog 
Devices Inc, 1980. 

3 Williams, Jim, "Signal Sources, Conditioners and Power 
Circuitry," Linear Technology Corp, Application Note 98, No- 
vember 2004. 

AUTHORS' BIOGRAPHIES 

Jim Williams is a staff scientist at Linear Technology Corp (www. 
linear.com) , where he specializes in analog-circuit and instrumen- 
tation design. He has served in similar capacities at National Semi- 
conductor, Arthur D Little, and the Instrumentation Laboratory 
at the Massachusetts Institute of Technology (Cambridge, MA). 
A former student at Wayne State University (Detroit), Williams 
enjoys sports cars, art, collecting antique scientific instruments, 
sculpture, and restoring old Tektronix oscilloscopes. 



ATEDN.COM [> 



[+1 Go to www.edn. 
com/ms4255 and 
click on Feedback 
Loop to post a com- 
ment on this articie. 



Mark Thoren is an applications manager for mixed- 
signal products at Linear Technology. He has degrees 
in agricultural mechanical engineering and electrical 
engineering — both from the University of Maine. A 
lifelong electronic and mechanical hobbyist, Thoren 
enjoys Silicon Valley's numerous swap meets and junk 
stores. When hes not in the valley itself, you can find 
him running in the hills surrounding it. 



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DESIGN 
NOTES 



Programmable Baseband Filter for Software-Defined 
UHF RFID Readers 

Design Note 432 
Philip Karantzalis 



Introduction 

Radio frequency identification (RFID) is an auto-ID tech- 
nology that identifiesanyobjectthatcontainsacoded tag. 
A UHF RFID system consists of a reader (or interrogator) 
that transmits information to a tag by modulating an RF 
signal in the 860MHz-960MHz frequency range. Typi- 
cally, the tag is passive— it receives all of its operating 
energy from a reader that transmits a continuous-wave 
(CW) RF signal. A tag responds by modulating the reflec- 
tion coefficient of its antenna, thereby backscattering an 
information signal to the reader. 

Tag signal detection requires measuring the time interval 
between signal transitions (a data "1 " symbol has a longer 
interval than a data "0" symbol). The reader initiates a tag 
inventory by sending a signal that instructs a tag to set 
its backscatter data rate and encoding. RFID readers can 
operate in a noisy RF environment where many readers 
are in close proximity. The three operating modes, single- 
interrogator, multiple interrogator and dense-interroga- 
tor, define the spectral limits of reader and tag signals. 
Software programmability of the receiver provides an 
optimum balance of reliable multitag detection and high 
datathroughput.Theprogrammable reader containsahigh 
linearity direct conversion I and Q demodulator, low noise 
amplifiers, a dual baseband filter with variable gain and 
bandwidth and a dual analog-to-digital converter (ADC). 
The LTC6602 dual, matched, programmable bandpass 
filter can optimize high performance RFID readers. 

The LTC6602 Dual Bandpass Filter 

The LTC6602 features two identical filter channels with 
matched gain control and frequency-controlled lowpass, 
and highpass networks. The phase shift through each 
channel is matched to ±1 degree. A clockf requency, either 
internal or external, positions the pass band of the filter 
at the required frequency spectrum. 

The lowpass and highpass corner frequencies, as well 
as, the filter bandwidth are set by division ratios of the 



clock frequency. The lowpass division ratio options are 
100, 300 and 600 and the highpass division ratios are, 
1 000, 2000, 6000. Figure 1 shows a typical filter response 
with a 90MHz internal clock and the division ratios set to 
6000 and 600 for the highpass and lowpass, respectively. 
A sharp 4th order elliptical stopband response helps 
eliminate out-of-band noise. Controlling the baseband 
bandwidth permits software definition of the operating 
mode of the RFID receiver as it adapts to the operating 
environment. 

An Adaptable Baseband Filter for an RFID Reader 

Figure 2 shows a simple LTC6602-based filter circuit that 
uses SPI serial control to vary the filter's gain and band- 
width to adapt to a complex set of data rates and encoding. 
(The backscatter linkf requency range is 40kHz to 640kHz 
and the data rate range is 5kbps to 640kbps.) 

For fine resolution positioning of the filter, the internal 
clock frequency is set by an 8-bit LTC2630 DAC. A OV 
to 3V DAC output range positions the clock frequency 
between 40MHz and 100MHz (234.4kHz per bit). The 
lowpass and highpass division ratios are set by serial 

ly, LT LTC and LTM are registered trademarks of Linear Technology Corporation. 
All other trademarks are the property of their respective owners. 




10 100 1000 

FREQUENCY (kHz) 

Figure 1. Filter Response for a 15kHz-150kHz Passband 



01/08/432 



SPI control of the LTC6602. The cutoff range for the 
highpass filter is 6.7kHz to 1 0OkHz and 667kH2 to 1 MHz 
for the lowpass filter. The optimum filter bandwidth set- 
ting can be adjusted by a software algorithm and is a 
function of the data clock, data rate and encoding. The 
filter bandwidth must be sufficiently narrow to maximize 
the dynamic range of the ADC input and wide enough to 
preserve signal transitions and pulse widths (the proper 
filter setting ensures reliable DSP tag signal detection). 

Figure 3 shows an example of the filter's time domain 
response to a typical tag symbol sequence (a "short" 
pulse interval followed by a "long" pulse interval). The 
lowpass cutoff frequency is set equal to the reciprocal of 
the shortest interval (fcuTOFF = l/IOps = 100kHz). If the 
lowpass cutoff frequency is tower, the signal transition 
and time interval will be distorted beyond recognition .The 
setting of the highpass cutoff frequency is more qualita- 
tive than specific. The highpass cutoff frequency must 
be lower than the reciprocal of the longest interval (for 



5V 



3V 



3V 



.0.1mF 



I CHANNEL ■ 
INPUT 



Q CHANNEL" 
INPUT 



TRANSMinER 
MUTE INPUT " 




LTC2630 
8-BIT DAG 



:o.imf 



Vddin Vdda Vood 



LTC6602 



CHANNEL 

— OUTPUT 

Q 

CHANNEL 

— OUTPUT 



DOUT 
NC 

^BIAS 



CLKIO 
PAR/SEl^ 
CLKCNTL 
SOD 

SCLK 
SDI 



VOCM GND 



-3V 



1 



FROM 
"ADC 

Vref 

OUTPUT 



SPI CONTROL OF LTC6602 SETS THE FILTER GAIN 
AND THE LOWPASS AND HIGHPASS DIVISION RATIO 



I I 

m scK SDI m 

Figure 2. An Adaptable RFID Baseband Filter 
with SPI Control 



Data Sheet Download i 



www.linearxom 



the example shown, highpass fcuTOFF < 1/20ps) and as 
high as possible to decrease the receiver's low frequency 
noise (of the baseband amplifier and the down-converted 
phase and amplitude noise). The lower half of Figure 3 
shows the filter's overall response (lowpass plus highpass 
filter). Comparing the filter outputs with a 10kHz and a 
30kHz highpass setting, the signal transitions and time 
intervals of the 10kHz output are adequate for detecting 
the symbol sequence (in an RFID environment, noise will 
besuperimposedontheoutputsignal).lngeneraUncreas- 
ing the lowpass fcuTOFF and/or decreasing the highpass 
f CUTOFF "enhances" signal transitions and intervals at the 
expense of increased filter output noise. 

Conclusion 

The LTC6602 dual bandpass filter is a programmable 
baseband filter for high performance UHF RFID readers. 
Using the LTC6602 under software control provides the 
ability to operate at high data rates with a single inter- 
rogator or with optimum tag signal detection in a multiple 
or dense interrogator physical setting. The LTC6602 is 
a very compact IC in a 4mm x 4mm QFN package and is 
programmable with parallel or serial control. 

References: 

1. The RFin RFID, Daniel M. Dobkin, 9/07, Elsevier Inc. 

2. Class-1 Generation-2 UHF RFID Protocol for Com- 
munications at 860 MHz to 960 MHz, Version 1.1.0, 
www.epcglobalinc.org/standards/specs/ 



TYPICAL TAG 
SYMBOL SEQUENCE 



LTC6602 100kH2 LOWPASS 















\ 




7 






7 









LTC6602 lOOkHz LOWPASS + HIGHPASS FILTER 




lOps/OIV 



Figure 3. Filter Transient Response to a Tag 
Symbol Sequence 



For applications help, 
call (408) 432-1900, Ext. 2020 



Linear Technology Corporation 

1630 McCarthy Blvd., Milpitas, CA 95035-7417 
(408)432-1900 • FAX: (408) 434-0507 • www.linearcom 



dn432 LT/TP 0108 241 K • PRINTED IN THE USA 

rrunm 

.^^W te(.:hnology 

© UNEAR TECHNOLOGY CORPORATION 2007 



EDITED BY CHARLES H SMALL 
AND FRAN GRANVILLE 



READERS SOLVE DESIGN PROBLEMS 



Use the MCLR pin as an output 
with PIC microcontrollers 

Antonio Munoz, Laboratorios Avanzados de Investigacion, 

Huesca, Spain, and Pilar Molina, Universidad de Zaragoza, Zaragoza, Spain 



Although microcontroller man- 
ufacturers try to offer design- 
ers products that almost exactly fit the 
needs of their designs, another output 
pin is often necessary. This situation 
is particularly true in small designs us- 
ing microcontrollers with eight pins 
or fewer. This Design Idea employs 
the Microchip (www.microchipxom) 
PIC10F222. The PIC10F222 comes in 
an SOT23-6 package and offers three 
I/O pins, one input pin, RAM, flash, 
and an ADC module. You must pro- 
gram these tiny microcontrollers, just 
as you do with their big brothers. To 
program these microcontrollers, you 
need the MCLR, two I/O pins (data 
and clock), and supply pins (V^^^^ and 
GND). To enter programming mode, 
you need MCLR and supply. Because 
the microcontroller must differenti- 



ate between normal and programming 
mode, the MCLR pin usually reaches 
a voltage of approximately 12V to en- 
ter programming mode. Thereafter, in 
normal operation, you can configure 
the MCLR pin either as an external re- 
set or as an input-only pin. 

This design uses one pin for analog 
input and the other three as outputs. 
The design thus requires an addition- 
al output. For that reason, this circuit 
uses the MCLR pin as an output. For 
simplicity. Figure 1 shows only the 
GP3/MCLR output circuit. To allow 
the GP3/MCLR pin to act as an out- 
put, the circuit uses the configurable 
weak puUups that this microcontroller 
offers. The selected function for the 
GP3/MCLR pin is input, and you must 
enable the global weak-puUup bit in 
the microcontroller's configuration 



100 nF 



^cc 
Q 



GPO/ANO 



GP3/MCLR 



GND PIC10F222 VDD 



GP1/AN1 



GP2/T0CK1 



^cc 

o 



1M 



1k 



LED 




Figure 1 Adding a MOSFET and associated circuitry to a PIC microcon- 
troller's MCLR input pin transforms the pin into an output. 



DIs Inside 

66 High-speed clannp functions 
as pulse-forming circuit 

68 Depletion-mode MOSFET 
kick-starts power supply 

70 Simple continuity tester fits 
into shirt pocket 

72 White LED shines 

from piezoelectric-oscillator supply 

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word- Although you cannot individu- 
ally configure weak pullups, this inabil- 
ity is not a problem because you con- 
figure all other pins as analog inputs or 
digital outputs- 

The weak pullups have a resistance 
of 20 to 150 kn, depending on supply 
voltages, so this circuit uses transistor 
to drive higher loads, such as the 
depicted LED. drives the transistor 
off when you deactivate the pullups. 
Because the transistor's gate is resis- 
tance-driven, the maximum toggle fre- 
quency depends on the chosen tran- 
sistor. The worst-case scenario occurs 
when you need to switch off Q^. and 
Q^'s gate-to-source capacitance deter- 
mine the transistor's switch-off time. 

Programming voltages for the MCLR 
pin are about 12V. Therefore, must 
withstand a gate-to-source voltage 
higher than this value. This design uses 
a MOSFET having a ± 18V withstand 
voltage. For this reason, you should not 
use digital MOSFETs. You can use this 
circuit with other PIC microcontrollers 
and with most RS08KA family micro- 
controllers from Freescale.EDN 



JANUARY 10, 2008 | EDN 65 



designideas 



High-speed clamp functions 
as pulse-forming circuit 

Marian Stofka, Slovak University of Technology, Bratislava, Slovakia 



Amplifiers with positive feed- 
back are the bases of signal- 
grade pulse-forming circuits- This 
setup ensures a triggerlike operation 
in which the input signal crosses the 
input- threshold level; in most cases, 
the input signal is a voltage signal- The 
most well-known of these triggers is the 
Schmitt trigger, which, by the way, will 
this year celebrate its 70th birthday. 
British scientist OH Schmitt in 1938 
originated the Schmitt trigger in the 
form of a two-stage amplifier with cur- 
rent feedback. The two active devices 
were vacuum tubes. 

The operation of a Schmitt trigger 
has the advantage of fast, almost-con- 
stant transition times of the output re- 
gardless of the slope of the input signal. 
One consequence of this type of opera- 
tion is the hysteresis in the I/O char- 
acteristic. In other words, the thresh- 



old shifts to a higher value before the 
positive-output transition, and it shifts 
to a lower value after switching to the 
positive-output level. You can set the 
amount of hysteresis — from zero to 
latch-up — for Schmitt-trigger circuits 
comprising discrete parts. Schmitt 
circuits find wide use in logic ICs, in 
which the hysteresis is rather high and 
fixed. 

As an alternative, you can use a cir- 
cuit — a fast-response voltage limiter, 
or clamper — as a pulse-forming cir- 
cuit. The input-voltage range is nar- 
rower than that of Schmitt-trigger cir- 
cuits, because, at low input voltages, 
the voltage limitation becomes inac- 
tive, and the circuit operates as a lin- 
ear amplifier. On the other hand, be- 
cause of its nonhysteretic behavior, the 
decision threshold of the input voltage 
is precise and equal for both directions 



AD8045 



HSMS-282L 




OOUT 



C| 

-1.2 pF 


avw-l- 

1 1 


M 

>^ 1 



NOTES: 

ALL COMPONENTS ARE SMD. 

ALL EXPERIMENTS WERE PERFORMED 

FOR ASYMMETRIC CONFIGURATION WITH D4 

SHORTED AND Rj^ OMITTED. 

FOR SYMMETRIC CONFIGURATION, USE THE IC2 

HAVING A SUFFIX "R" (RING-OUAD DIODES); 

EVENTUALLY, USE "P" (BRIDGE-OUAD DIODES) INSTEAD 

OF "L" AND COMPLETE THE CIRCUIT WITH D4 

AND Rt2 as DEPICTED. 



Figure 1 This clamping circuit uses diodes to achieve nonlinear feedback. The 
circuit employs a single diode in one feedback path and two diodes in the 
other. The dual-diode configuration offers cleaner switching. 



of output-level transitions. Figure 1 
shows one example of such as circuit. 
The voltage limiter in Figure 1 is an 
inverting amplifier with a highly non- 
linear negative feedback. For output 
voltages ranging from —0.3 to +0.6V, 
the feedback impedance is high be- 
cause each of the diodes is noncon- 
ducting. The forward-voltage drop of 
the selected Schottky-barrier diodes 
determines these voltage limits (Ref- 
erence 1 ) . The voltage gain of the in- 
verting amplifier is thus almost that of 
the op amp's open-loop gain. 

Whenever the output voltage ex- 
ceeds these limits, diode D^, or 

— depending on the polarity of the 
output voltage — starts to conduct. 
The differential gain of the amplifier 
then drops to the value of — Rj/ZR^^ 
and — Rj/Rj^, respectively, where 
is the equivalent-series resistance of 
a single diode. The action clamps the 
output voltage to approximately 0.8V 
and to —0.4V even for large input 
voltages. The figure uses an Analog 
Devices (www.analog.com) AD8045 
VHSIC (very-high-speed integrated- 
circuit) op amp because its slew rate 
exceeds the value of IV/nsec (Refer- 
ence 2). 

Figure 1 's circuit has an asymmetri- 
cal-limiting configuration to compare 
the single feedback diode with two se- 
ries-connected diodes having a trans- 
verse resistor, R^^ between their mid- 
points and ground. The clamping cir- 
cuitry comprising 0^,0^, and R^^ offers 
higher off- isolation between the out- 
put and the input of the op amp than 
that of the single diode, D^. When 
is on, you can observe small, weakly 
damped oscillations at approximately 
200 MHz in the output waveform. Os- 
cillations manifest themselves less at 
the beginning of turn-on of the and 
diodes.EDN 

REFERENCES 

a "Surface Mount RF Schottky Bar- 
rier Diodes," Avago Technologies, 
www.avagotech.com/. 
a "AD8045 3 nV/VHz Ultralow 
Distortion Voltage Feedback High 
Speed Amplifier," Analog Devices, 
2004, www.analog.com/en/prod/ 
0„ad8045,00.html. 



66 EDN I JANUARY 10, 2008 



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INTEGRATE AUDIO AMP 
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CONTROL _ 



DIFFERENTIAL 
INPUTS 



Hh 



NEGATIVE 
CHARGE 
PUMP 



INTERFACE 
+ 

CONTROL 



DYNAMIC 
CURRENT 
REGULATORS 



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MAX8678 



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OUTPUT 
0,1mA TO 24mA 



Maxim's SOLUTION = 9mm2 
50% SMALLER 
THAN COMPETITION 



START 




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COMPETITIVE SOLUTION = 18mm2 

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CELL PHONE 
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Down to 0.1mA 

♦ Low 140pA Quiescent Current 

♦ Single-Wire, Serial Pulse Dimming Interface 



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♦ Single-Supply Operation 

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♦ -9dB to +18dB Gain Settings in 3dB Steps 

♦ Integrated Click-and-Pop Suppression 

♦ No Output-Coupling Capacitors, Snubber 
Networks, or Bootstrap Capacitors Required 



t2500-up recommended resale. Prices provided are for design guidance and are FOB USA. International prices will differ due to local duties, taxes, and exchange rates. Not all pack- 
ages are offered in 1k increments, and some may require minimum order quantities. 



www.maxim-ic.com/MAX8678-info 
FRS Audio Design emSe—Sent Within 24 Hours! 

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electronics marketing 

1-800-332-8638 



Distributed by IVIaxim Direct, Avrnet Electrornics IVIarketirng, Digi-Key, arnd Newark. 
The IVIaxim logo is a registered trademark of Maxim Irntegrated Products, he. 
© 2008 Maxim Integrated Products, he. All rights reserved. 



designideas 



Depletion-mode MOSFET 
kick-starts power supply 

Gregory Mirsky Milavia International, Buffalo Grove, IL 



MH Many switch-mode power sup- 

■H plies use "kick-start" circuits 
to initialize their offline operation. 
These circuits may be simple resistors, 
such as International Rectifier's (www- 
irfxom) IRIS4015, or more compli- 
cated arrangements built with bipo- 
lar transistors or MOSFETs (Refer- 
ence 1). These transistors provide the 
initial current for the flyback or PFC 
(power-factor-correction) IC. When 
such a power supply starts operating 
in normal mode, a supply voltage from 
a dedicated winding keeps supplying 
the PFC IC, thus reducing power con- 
sumption of the kick-start circuitry. 

Such schemes reduce — but do not 
eliminate — the power consumption of 
the kick-start circuitry, because the ac- 
tive component is usually a high- volt- 
age bipolar transistor or high-voltage 
enhancement-mode MOSFET These 
transistors' base or gate requires for- 
ward-biasing with respect to the emit- 
ter or the source for normal operation. 
Therefore, a power loss always occurs 
in the circuits that keep the transis- 
tors in the off state. Unfortunately, 



engineers pay too little attention to 
depletion-mode MOSFETs, which re- 
quire no forward-biasing for normal 
operation and, moreover, require gate 
potentials below the source. These 
valuable properties of depletion-mode 
MOSFETs suit them for a role in no- 
loss kick-start circuits for power sup- 
plies. 

Figure 1 shows a conventional PFC 
circuit whose IC initially receives pow- 
er from the output through a deple- 
tion-mode MOSFET, Q^, a DN2470 
from Supertex (www.supertex.com. 
Reference 2). Q^'s source feeds PFC 
IC^ with an initial supply current of 
approximately 10 to 15 mA or less 
depending on the IC model. A brief 
power dissipation of approximately 4 
to 6W can do no harm to the MOS- 
FET soldered to a copper pour. If you 
have concerns about the MOSFET's 
health, you can use an IXTY02N50D 
from Ixys (www.ixys.com. Reference 
3). Resistors R^ and R^ set up Q^'s 
working point to obtain the minimum 
required current. Zener diode lim- 
its voltage across IC^ to approximate- 



ly 15V for an input voltage of 18V, 
which is usually necessary for most 
PFC ICs and is less than the maxi- 
mum for MOSFET Q^. 

When IC^ starts working normal- 
ly, the secondary winding of the PFC 
inductor, L, generates the IC's supply 
voltage, which diodes and and 
capacitors C^ and C^ condition. Tran- 
sistor Q2 keeps feeding zener diode 
and IC^ for a short interval. Eventu- 
ally, bipolar transistor gets its base 
supply through resistor R^ from diode 
D^, turning on and clamping Q^'s gate 
to ground. Q^'s power source is the 
IC's positive-supply potential of ap- 
proximately 15 V, which is more than 
enough to shut off Q^. The residual 
thermal current of 10 to 20 |jlA pro- 
duces no substantial power loss.EDN 

REFERENCES 

a "IRIS4015(K) Integrated Switcher," 
Data Sheet No. PD60190-C, Interna- 
tional Rectifier, www.irf.com/product- 
i nf o/datasheets/data/i ris4 1 5 . pdf . 
a "DN2470 N-Channel Depletion- 
Mode Vertical DMOS PET," Supertex 
Inc, www.supertex.com/pdf/ 
datasheets/DN2470.pdf. 
a "High Voltage MOSFET: IXTP 
02N50D, IXTU 02N50D, IXTY 
02N50D," Ixys, http://ixdev.ixys.conn/ 
DataSheet/98861.pdf. 







RECTIFIER 

— M— 

GBU-608 


LINE 
FILTER 













FDLL4148 



D2 : 

FDLL4148 



1 |xF " 



O2 
=20 ^JLF 
35V 

R5 
100k 



100k 



D3 

FDLL4148 



^cc 
IC, 

PFC GATE 
CONTROLLER 

GND 



_L C3 
"p.1 ^JLF 



CSD06060G 




R3 
1M 



CMPT2222A 



.BZX84C15- 
LT1 



-O420V 



Q2 
DN2470 



L 



O4 
. 470 |xF 
450V 



SETS 
R4 INITIAL 
^■2M CURRENT 



Figure 1 A depletion-mode, high-voltage MOSFET provides a kick-start for a PFC 10. During normal operation, the MOSFET 
switches off and dissipates negligible power. 



68 EDN I JANUARY 10, 2008 



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AUTOMOTIVE IC 
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All reliability reports are readily available on our website. 

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Tine Maxim logo is a registered trademark of Maxim Integrated Products, Inc. © 2008 Maxim Integrated Products, Inc. All rights reserved. 



designideas 



Simple continuity tester 
fits into shirt pocket 



Tom Wason, Phoenixville, PA 

MH This Design Idea describes a 
mtM handy continuity tester with 
two modes of operation: It may sound 
if it detects continuity between its two 
probes, or it may sound when it de- 
tects no continuity. The second option 
permits testing for intermittent cable 
breaks. Response must be sufficiently 
fast to permit swiping a probe across 
perhaps 100 pins to instantly find a 
connected pin. The tester may also 
identify microfarad or larger capaci- 
tance between two conductors. 

To properly test for continuity, the 
tester's voltage and current are limited 
so that low-power semiconductors do 
not suffer overstress or appear as a con- 
nection between two conductors. The 
tester must protect itself if you acciden- 
tally connect it across an energized cir- 
cuit or a charged capacitor. Power con- 
sumption must be low so that if you 
accidentally leave the tester on over- 
night, it will not discharge the battery. 
The tester must operate even with low 
battery voltage. 

Continuity requires a threshold of 
less than 20011. Depending on bat- 
tery voltage, that threshold may even 



be son. The tester's open-circuit volt- 
age is less than 0.5V. Its short-circuit 
current is approximately 1 mA. Val- 
ues are low so that the tester doesn't 
mistake a Schottky-barrier rectifier for 
continuity. When the tester is silent, it 
draws slightly more than 1 mA of cur- 
rent from a 9V battery. You can con- 
nect the probes for a few seconds across 
any voltage from —50 to +200V with- 
out damage. 

A feedback circuit comprising 
PNP and NPN transistors main- 
tains voltage on the gate of IGFET 
(insulated-gate field-effect transistor) 
Q3 at less than 1.4V despite a 680-kn 
puUup resistor, R^, and current from 
D^ (Figure 1). When you short the 
probes, you divert more base cur- 
rent to the probes, and less current 
flows through D^. Eventually, can 
no longer maintain a low gate volt- 
age. As the gate voltage exceeds 1.8V, 
Q^'s drain-to-source current causes 
to become nonconductive. A l-MH 
puUup resistor, R^, then applies 9V to 
Q^'s gate, causing the tester to sound, 
announcing continuity. 

Without a conducting collector. 



— 9V 

1 



Ri 

5.6k 




PROBE 



BLACK 



R4 
680k 



R5 
1M 



Re 
1M 



Di ^-^N5089 
'1N5819 



Q3 
BS170 



C2 S SOUND 
"p.01 |jlF 



m 

^ ■'^R 




: Ci 
0.1 |jlF 



1 N4003 



D4 
1 N4003 



Figure 1 This simple continuity tester is switch-selectable to sound on either shorts 
or opens. It prevents a user from accidentally connecting it across live circuits. 



i 



Q^'s gate voltage approaches 9V. Cur- 
rent would then leak through Q^'s col- 
lector-to-base path. Diode D^ blocks 
Q^'s gate voltage from leaking to the 
shorted probes. 

The tester detects instantaneous con- 
tinuity even when you quickly swipe a 
probe across 100 pins. Capacitor C^ and 
puUup resistor R^ extend Q^'s low gate- 
voltage response by 20 msec. Thus, the 
tester sounds slightly longer to indicate 
that it has established connectivity and 
does not miss a conductive pin during 
a fast swipe. 

Probe current charging a capacitor 
may also create a short beep. The 20- 
msec extended beep means that the 
tester detects even lO-fxF or smaller 
capacitors. With practice, you can esti- 
mate capacitance within decades from 
the beep's period. 

Diodes D^ through D^ block destruc- 
tive currents if probes touch an ener- 
gized circuit. Resistor R^ must be at 
least ViW to withstand current from 
an energized circuit for a few seconds 
without damage. 

To test for cable continuity, the tes- 
ter sounds only during a broken con- 
nection. In this case, firmly connect 
the probes to both ends of the cable. 
Switching changes the tester's func- 
tion so that drives the buzzer during 
a cable break. 

You can modify the circuit to be a 
better cable tester by reducing the value 
of resistor R^ to 4-7 kll and omitting 
capacitor C^. With these modifications, 
detecting loss of continuity occurs at a 
threshold resistance of less than lOOH. 

Unfortunately, a continuity tester 
may create noise currents that feed 
back into the sensitive QJQj detec- 
tor. Three circuit features minimize 
that noise. First, capacitor C^ connects 
across the buzzer. Second, IGFET 
acts as a buffer. Last, diode D^ grounds 
and separately from ground for 
Q.andQ^. 

The circuit performs even when a 
battery voltage is less than 6.5V. How- 
ever, lower battery voltage means that 
the tester detects continuity at a high- 
er threshold resistance. You may install 
the entire tester in a plastic case small- 
er than a pack of cigarettes.EDN 



70 EDN I JANUARY 10, 2008 



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electronics mariceting 

1-800-332-8638 



Distributed by IVIaxim Direct, Avrnet Electrornics IVIarketirng, Digi-Key, arnd Newark. 
The IVIaxim logo is a registered trademark of Maxim htegrated Products, he. 
© 2008 Maxim htegrated Products, he. All rights reserved. 




designideas 



White LED shines 

from piezoelectric-oscillator supply 



T, 



TA Babu, Chennai, India 

LED drivers that receive their 
power from a single cell are re- 
ceiving a great deal of attention. To 
generate the high voltage for illumi- 



-01.5V 



^100 |xH 



PIEZOELECTRIC 
CERAMIC 
BUZZER 




Figure 1 A piezoelectric ceramic buzzer serves as the 
oscillator in this flyback supply that lights a white LED 
using a single cell (not shown). 



nating a white LED from a low-volt- 
age power supply basically requires 
some form of an electronic oscillator, 
and one of the simplest is a piezoelec- 
trie buzzer. An un- 
usual application of 
a piezoelectric trans- 
ducer serves as an os- 
cillator and drives a 
white LED (Figure 
1). The piezoelec- 
tric diaphragm, or 
bender plate, com- 
prises a piezoelectric 
ceramic plate, with 
electrodes on both 
sides, attached to a 
metal plate made of 
brass, stainless steel, 
or a similar material 
with conductive ad- 



-WHITE LED 



hesive. The circuit uses a three-ter- 
minal piezoelectric transducer. In this 
transducer, the diaphragm has a feed- 
back tab on one of its electrodes. The 
oscillation is a result of the resonance 
between the inductor and the element, 
which is capacitive. The frequency of 
operation is: 1/(27tVLC), where 
L is the value of the inductor and C 
is the capacitance of the piezoelectric 
element. 

With the initial application of po- 
tential to the circuit in Figure 1 , tran- 
sistor turns on. When the transistor 
conducts, the current through inductor 
L^ increases gradually, and the poten- 
tial across the plates flexes the piezo- 
electric ceramic. This flexing generates 
a negative potential at the feedback 
tab, which feeds back to the base of the 
transistor. The transistor then switches 
off. When turn-off occurs, the stored 
energy in the inductor dumps into the 
LED. This flyback voltage is sufficient 
to light the LED.EDN 



Ultra-Low Dropout Regulators 



+2.5 to +5.5V 



Battery 
Input 



ViN 

■_L 



GND 

■_L 



SHDNM 




1 


IVOUT 


m 




AS1353 




1 


BP 



+1.5 to 3.6V 



Preset 
Output 



► Ultra-Low Noise (9|jVrms, 92dB PSRR) 

► High Voltage up to 20V 

► High Accuracy of 0.5% 

► Low Power Consumption 

► *User-Programmable Output Voltages 



I 



samples online at »f 



Part No. "^^^H 


AS1351 


AS1357 AS1352 


AS1353 


AS1356 


AS1358 


AS1359 


AS1361 AS1362 


AS1360 


AS13985 


AS13g86 


Outputs 


2 


3 4 


1 


1 


1 


1 


1 1 


1 


1 


2 


Dropout Voltage (mV) 


200 


200 200 


60 


60 


70 


140 


70 140 


400 


45 


45 


Output Current (mA) 


200 


200 200 


150 


150 


150 


300 


150 300 


250 


150 


150 


Feature 


OTP* 


OTP* OTP* 


Low Noise 


Power-OK 


Low Noise 


Low Noise 


Low Noise + POK Low Noise + POK 


High Voltage 


WL-CSP 


WL-CSP 


Supply Current (pA) 


125 


175 225 


70 


70 


40 


40 


40 40 


1.5 


95 


135 



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West Coast (408) 345-1790 East Coast (919) 676-5292 
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72 EDN 



JANUARY 10, 2008 



3 GSPS 8-Bit ADC with Over 3 GHz 
Full Power Bandwidth 




1.9W Ultra Low Power ADC from the PowerWise® Family is Easily Interleaved for 
6 GSPS Operation 



ADC083000 Features 

• 1.9W operating power consumption is less than half 
competitive solutions 

• 3 GHz full power bandwidth 

• Bit Error Rate (BER):10-is 

• Single supply operation: +1.9V 

• Integrated 1:4 output demultiplexer 

• Adjustable input full-scale range and offset 

• Guaranteed no missing codes 

• Integrated 4K capture buffer available (ADC08B3000) 

ADC083000 Benefits 

• Clock phase adjust for multiple ADC synchronization 
allows 6 GSPS operation with 2 interleaved ADCs 

• Test pattern simplifies high-speed data capture 

• Serial interface for controlling extended functionality 

• Reference board available with LMX2531 clock 
conditioner and LMH6555 high-speed amplifier, 
for inputs between DC and 750 MHz 



Applications 

Communications infrastructure, test and measurement 
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For reference design, samples, datasheets and more, visit: 



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Na tional 

Semiconductor 

The Sight & Sound of Information 



National Semiconductor Corporation, 2007. National Semiconductor, PowerWise, and fjiare registered trademarks of National Semiconductor Corporation. All rights reserved. 



EDITED BY SUZANNE DEFFREE | | | 

supplychain 



LINKING DESIGN AND RESOURCES 



2007: a less-than-memorable year for DRAM 



If there were ever proof that 
optimism doesn't necessari- 
ly lead to success, the 2007 
DRAM market is such proof. 
Microsoft's Vista spurred 
hopes of increased demand 
for 1-Gbyte DRAM, but those 
hopes went unfulfilled, even 
though vendors ramped up 
capacity in preparation. "A lot 
of it had to do with how 2006 
ended," says Paul Zecher, com- 
modity manager for memory at 
independent distributor Con- 
verge (www.converge.com). 
"There was so much capacity 
coming onboard ... that, once 
we got into the holidays and 
consumption started to slow 
down, those guys were up 
and running. A lot of it had to 
do with their expectation that 
Vista would take off going in- 
to the first half of '07, as well. 
They were banking on that to 

MlJJJilllJ.IVik 



happen, and it didn't." 

According to Zecher (photo), 
1 -Gbyte DRAM is the "module 
of choice" for Vista PCs, and, as 
such, vendors heavily ramped 
up production. Overcapac- 
ity led to deteriorating market 
prices, many of which fell be- 
low cash costs in the final quar- 
ter of 2007 That oversupply is 
expected to bleed into 2008's 
first-and, possibly, second- 
quarter. "The DRAM recovery 
will be driven by the supply 
side, with inventory dwindling 
and growth in bit production 
decreasing," says Nam Hyung 
Kim, director and chief ana- 



lyst for memory ICs and stor- 
age systems at iSuppli (www. 
isuppli.com). "This [decrease] 
will cause availability to tight- 
en and prices to rise. Howev- 
er, this scenario assumes that 
suppliers' behavior will be ra- 
tional, and they will not engage 
in any massive production in- 
creases that could send DRAM 
pricing into a new dive." 

Zecher also notes the possi- 
bility of corporate upgrades fi- 
nally adjusting to Vista systems 
in 2008 but warns that this ad- 
justment depends heavily on 
global economies (see "De- 
mand-driven downturn pos- 
sible in 2008," right). "If com- 
panies are hurling or there are 
layoffs, more than likely they 
are not going to start upgrad- 
ing their systems. And Vista is 
not as easy as a lot of the other 
Microsoft upgrades," he says. 




CALIFORNIA VETOES ROHS-EXPANSION BILL 

California Governor 

Arnold Schwarzenegger 
has vetoed an assembly 
bill that would have more 
closely aligned California's 
ROHS (restriction-of-hazardous-substances) 
law and regulations with the EU (European 
Union) ROHS directive. The bill, AB 48, would 
have expanded California ROHS to include all 
electrical and electronic equipment, as opposed 
to its current requirements for "covered elec- 
tronic devices," which include nine video-display 
devices that the state's Department of Toxic 
Substances Control (www.dtsc.ca.gov) regula- 
tions list. The bill also aimed to require that all 
electrical and electronic equipment that manu- 




facturers sell in California as of Jan 1 , 201 0, 
comply with EU ROHS stipulations for lead, 
mercury, cadmium, and hexavalent cadmium. 

Schwarzenegger sent the bill back to the 
California legislature in the fall, stating in a | 
memo that the bill's approach "is largely un- 
workable and instead of the benefits it seeks to 
accomplish, could ultimately result in unintend- 
ed and potentially more harmful consequences." 

The governor noted the exemption language 
for spare parts and refurbished products, claim- 
ing that, as written, the bill would make many 
electronic products prematurely obsolete and 
force their retirement years earlier than neces- 
sary. The California legislature is expected to try 
to pass this or a similar bill again in 2008. 



DEMAND-DRIVEN < 
DOWNTURN i 
POSSIBLE IN 2008 ! 

The new year could bring / 
a demand-driven downturn, 
Gartner Inc (www.gartner. 
com) cautions. Government 
economists and financial 
counselors have warned that 
the United States could ex- 
perience a recession due to 
subprime-rate-mortgage debt 
and the US-initiated, and now 
global, "credit crunch." As a 
result, Gartner believes the 
semiconductor industry could 
see this downturn in 2008. 

Although the effects of 
such a downturn would be 
milder than those of the 2001 
bubble burst, they depend on 
the electronics segment in the 
supply chain, Gartner says. 
The research company notes 
that the downside risk is lower 
for semiconductor companies, 
given a slow 2007, and higher 
for capital-equipment com- 
panies, given the substantial 
memory-related capacity 
investments in 2007 

According to Gartner, 
equipment companies' 
revenue in a mild recession 
could drop as much as 1 5 
to 20% in 2008, whereas 
chip-company revenue may 
decline in the mid-single-digit 
range. Gartner predicts that, 
if a recession were to hit the 
United States, the possible 
semiconductor-industry 
down cycle would likely 
extend into 2009, depending 
on the impact on other 
global economies and the 
severity and speed of the US 
recovery. 



EDN I JANUARY 10, 2008 



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productroundup 



DISCRETE SEMICONDUCTORS 





Power MOSFET 
suits ORing applications 

Targeting MOSFET ORing applications, the 
STV300NH02L power MOSFET claims to 
increase system reliability in paralleling-configu- 
ration power-server applications- The 20V device 
provides a typical O-S-mll on-resistance. Available 
in a PowerSO40 package, the STV300NH02L 
MOSFET costs $4.50 (1000). 
STMicroelectronics, www.st.com 

N-channel MOSFETs 
target mobile electronics 

Aiming at dc/dc-converter applications for 
mobile electronics, the 30V OptiMOS3 M se- 
ries N'channel-MOSFET family targets applications 
with a 5V drive. The BSC100N03MS G with a lO-mll on^-re^ 
sistance provides an 11 -nC maximum gate-charge rating. The 
MOSFETs are available in a 30-mm^ SS08 (SuperSOS) pack- 
age or in a 3X3-mm S308 (shrink SuperSOS) package. The 
SS08 achieves a 2-mn on-resistance at 4-5 V and 1.6-mH on- 
resistance at lOV, and the S308 achieves a 43 -mil on-resis- 
tance at 4-5 V and 3. 5 -mil on-resistance at lOV. The SS08 
OptiMOS3 M series costs 88 cents for the 1.6-mH device; the 
S308 packaging costs 56 cents for the 3. 5 -mil device. 
Infineon Technologies, www.infineon.com 

Switching MOSFETs 
provide higher drain currents 
for synchronous dc/dc converters 

Extending the vendor's UMOS V-H high-speed 
switching-MOSFET series, the TPCA8012-H and the 
TPCA8019-H N-channel MOSFETs suit synchronous dc/dc 
converters in power supplies for servers, desktop and mobile 
computers, and portable- electronics devices. The TPCA80 12- 
H features a 40 A maximum drain current, a maximum 30V 
drain-to-source voltage, and a 4-9-mn on-resistance with a 
lOV maximum gate-to-source voltage. The TPCA8019-H 
provides a 45 A maximum drain current, a maximum 30V 
drain-to-source voltage, and a 3.1 -mil on-resistance with a 
lOV maximum gate-to-source voltage. Available in SOP- Ad- 
vanced packages with a 5x6-mm footprint, the TPCA8012- 
H and TPCA8019-H cost 45 and 60 cents, respectively. 
Toshiba America Electronic Components, www.toshiba. 
com/taec 



76 EDN I JANUARY 10, 2008 



You need this dock generator 

CC635 - Precise, low jitter cloclcs from DC to 2.05 GHz 




CG635...$2490 (u.s. ust) 



• Square wave clocks from DC to 2.05 GHz 

• Random jitter <1 ps (rms) 

• 80 ps rise and fall times 

• 16-digit frequency resolution 

. CMOS, LVDS, ECL, PECL, RS-485 

• Phase adjustment and time modulation 

The CG635 Synthesized Clock Generator provides square wave clocks 
between DC and 2.05 GHz that are clean, fast and accurate. With jitter 
less than 1 ps, transition times of 80 ps, and 16 digits of frequency 
resolution, the CG635 will meet your most critical clock requirements. 

The instrument can provide clocks at virtually any logic level via coax 
or twisted pairs. The outputs have less jitter than any pulse generator 
you can buy, with phase noise that rivals RF synthesizers costing ten 
times more. 

Optional OCXO and rubidium timebases improve frequency stability by 
lOOx and 10,000x over the standard crystal timebase. And an optional 
PRBS helps you evaluate high-speed serial data paths. 

Whether you are trying to lower the noise floor of an ADC, increase 
SFDR of a fast DAC, or squash the bit error rate in a SerDes, the CG635 
is the tool you need to get the job done. 




280.0 mV/div 



■500 ps/div 
E00.0 mV/div 



Clock and PRBS signals at 622.08MHz 

Plot shows complementary clock and PRBS (opt. 1) 
outputs at 622.08 Mb/s with LVDS levels. Traces have 
transition times of 8 Ops and jitter less than 1 ps (rms). 



-140 
-150 















































iJ — 622.0 


8 MHz clock 






















MHz clock 





















































10 100 Ik 10k 

Offset Frequency (Hz) 

Phase noise for 10 MHz and 622.08MHz outputs 



ft : REF B : REF 

10.00 -10.00 
I dBm ][ 




DIV DIV CENTER 100 000 000.000 

IB. 00 10.00 SPAN 100 000 000.000 

KBWr ] Kh,- '.-n :c'4.5 min RANGE : R- 10. T--10 
RBW= 1 KHZ 



RF Spectrum of a 100 MHz clock 

Graph shows a 100 MHz span around a 100 MHz clock. Only 
two features are present: the clock at 100 MHz, and the 
spectrum analyzer's noise floor (around -82 dBc). 



Stanford Research Systems^ Inc. 



Phone: (408)744-9040 
www.thinkSRS.com 



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ADVERTISER INDEX 



Company ^^^^^^^^H 


Page 


Actel Corp 


C-3 


Agilent Technologies 


27 


Analog Devices Inc 


12 




19 




21 


Attnel Corp 


2 


austriamicrosystems AG 


72 


Avnet Electronics Marketing 


14 




75 


CML Microcircuits (UK) Ltd 


76 


Coilcraft 


17 


Digi-Key Corp 


1 


Fairchild Semiconductor 


11 


Freescale Semiconductor 


23 


Intersil 


29 




51,57 


Kemet Electronics Corp 


55 


LeCroy Corp 


8 


Linear Technology Corp 


59 




61,62 




63-64 


Mathworks Inc 


53 


Maxim Integrated Products 


67 




69 




71 


Micrel Semiconductor 


C-4 


Microchip Technology 


43 


Molex Inc 


31 


Murata Power Solutions Inc 
(formerly C&D Technologies Inc) 


44 


National Instruments 


32 




39 


National Semiconductor 


4 




33-34 




73 


NEC Tokin Corp 


42 


Noritake Co Inc Elec Div 


24 


NXP Semiconductors 


41 


Pico Electronics 


30 




35 




40 


Power-One Inc 


49 


Rabbit Semiconductor 


13 


Radicom Research 


79 


Signal Consulting Inc 


60 


Stanford Research Systems Inc 


77 


Tern 


79 


Texas Instruments 


C-2 




3,6 


Toshiba America 


46 


Trilogy Design 


79 


Xilinx Inc 


45 


EDN provides this index as an additional service. 
The publisher assumes no liability for errors or 
omissions. 




p 



mart 



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and current products. 




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• Fax, Voice playback and recording 
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Easily create and manage multi-level 
parts lists and specs, calculate costs, 
generate shopping and kit lists, print 
labels, generate RFQs and PCs and 
mucli more... 



Parts 
Vendors 



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and Vendor Database 



Get the fun function DEMO at 

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productmart 

For innore information 
on how to place 
your ad 
in this section 
please contact: 



BARBARA COUCHOIS 

T: 781-734-8405 

E-mail: botoo,coucfiois@reedbusiness,com 



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Host USB, Ethernet, 



• 4.3x3.3", Easy to ''-^ 
program in C/C++ 

• Two USB Host Ports 
forUSBFIasiiDisi(,/ 
Keyboard/mouse 

• 1Q0MBase-T 
Etiiernet, 6 RS232, 
i/0,5.7"TFT 
support 

• 16 cl) 24-bit 
ADC,11ch 
12-bitADC, 
8ch. 16-bit DAC 

• CompactFlash card with FAT file system support 

• Solenoid drivers, and Reed Relays 



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JANUARY la 2008 



EDN 79 



EDITED BY RON WILSON 




CHART YOUR COURSE 



AHEAD 



TO APEC 2008 

The 2008 Applied Power Electronics Conference 
will happen February 24 to 28 in Austin, TX. Take 
your pick of controversies, tutorials, and papers on 
advanced topics in power-electronics design; they'll 
all be there. Highlights will include a number of ses- 
sions on digital control of power circuits, including 
theory modeling, and prototyping; on the quest for 
greater efficiency; on electromagnetic analysis for 
power-supply folks; and (inevitably) on power elec- 
tronics in alternative-energy applications. 



AROUND 



r 



AT THE IMPLICATIONS 
OF THE CREDIT CRISIS 

What does the global credit crisis that grew out 
of the US subprime-mortgage fiasco have to do 
with electronics? Unfortunately there are sev- 
eral close links. Directly any time less money is 
available to households— such as with the drying 
up of home-equity loans in the United States- 
demand falls for the big-ticket consumer items 
that carry a lot of electronics content. What's 
more pernicious is that the spreading credit 
crunch and conscious tightening of business 
credit in China are making credit scarcer for 
businesses as well, reducing capital spending on 
electronics. In particular, that most capital-inten- 
sive purchase-a production fab-may become 
impossible without smoothly functioning global 
credit markets. The situation could improve if a 
country were willing to step up and pay the bill 
on its own, perhaps using a sovereign-wealth 
fund to invest directly in capital equipment. But 
that's a step no one has taken yet. 



Bi 



ACK 



AT A COMPUTER-BASED 
COMBAT AUTOPILOT 

An airborne digital computer now in 
production can fly a jet aircraft through 
all phases of supersonic combat and is 
small enough to fit in the cabinet of a 
2 1 -in. table-model TV. The computer 
can make 9600 arithmetical computa- 
tions in one second and 6250 decisions 
in a minute. Combining information from 
ground-control stations and the aircraft's 
radar, the computer takes in 61 types of 
information and outputs 30 types, in the 
process monitoring 16 navigation and 
flight-control functions on a 1.8-second 
cycle. Instead of tubes, the computer 
contains 4000 diodes, and 75% of its 
wiring is etched circuitry 
—Electrical Design News, January 1 958 



80 EDN 



JANUARY 10, 2008 






Pprtables everywhere just got a lot cooler. 




150 mW 



100 mW 



50 mW 



mW 




ARM-enabled Ml IGLOO: The lowest power FPGA 
with the industry-standard 32-bit microprocessor. 

Extend portable battery life by 10x without compromising 
time-to-market or budgets. Actel M1 IGLOO Flash FPGAs 
save power in every mode of operation, and never need 
extra components that add bulk and cost. 



At 148 \i\N, IGLOO static power 
use is barely visible compared 
to equivalent gate parts from 
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more power. 




ARM 



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M1 IGLOO devices can save you a lot more than just power. Like all Actel 
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POWER MATTERS 



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CML/LVPECL/LVDS Buffers Prevent Metastable Conditions With Fail-Safe Input (FSI) 




Micrd'S new buffers, fanouts, multiplexers, and dividers 
include a special Fail-Safe Input (FSI) circuit that senses invalid 
or no input signal. When the input signal fails, the FSI latches 
the output to the last state and this will eliminate metastable 
conditions and guarantee a stable output. 



FSI is an ideal feature for Hot Swap applications in rack-based 
equipment. No metastable or indeterminate state will occur 
at the output under these conditions. For more information, 
contact your local Micrel sales representative or visit us at: 
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FSI Parts 


Description 


Fmax 


1K Price 


SY58603/4/5U New! 


1:1 CML/LVPECI_/LVDS Buffer 


4.25 Gbps 




$1.70 


SY58606/7/8U New! 


1 :2 CMI_/LVPECI_/LVDS Buffer 


4.25 Gbps 




$1.95 


SY58609/10/11U New! 


2:1 CMI_/LVPECI_/LVDS Buffer 


4.25 Gbps 




$1.95 


SY89835U New! 


1 :2 LVDS Fanout Buffer 


3.2 Gbps 




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SY89464/5U 


1:10LVPECLyLVDS Fanout w/2:1 IVlUX 


2 GHz 




$3.92 


SY89837/8U 


1 :8 LVPECiyLVDS Fanout w/ 2:1 MUX 


2 GHz 




$3.94 


SY89843/4U 


1 :2 LVPECULVDS Fanout w/ 2:1 MUX 


2 GHz 




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SY89228/9U New! 


^3 and ^5 LVPECI_/LVDS Divider w/ Internal Term 


1.5 GHz 




$1.85 


SY89230/1U New! 


-^3 and -i-5 LVPECI_/LVDS Divider w/ Internal Term 


3.2 GHz 




$2.69 


SY89467/8U New! 


1 :20 LVPECI_/LVDS Fanout w/ 2:1 MUX 


2 GHz 




$6.15 


SY89846/7U New! 


1 :5 LVPECULVDS Fanout w/ 2:1 MUX 


2 GHz 




$2.49 



miCilEL 

InnovaHon Through Technology™ 

www.miGrel.com 



© 2008 Micrel, Inc. All rights reserved. MLF is 
a registered trademarl< of Amkor Teclnnology. 



1,000 piece suggested resale unit price. FOB. USA.